Power amplifier with multiple power supplies

ABSTRACT

A power amplifier for receiving an input signal and providing a corresponding amplified output signal. One embodiment of the power amplifier includes a positive half circuit for supplying power to an amplifier during positive half waves of the output signal and a negative half circuit for supplying power to the amplifier during negative half waves of the output signal. Each half circuit has a main power supply, which is typically a switching regulator, and which supplies a first power signal to the amplifier. The slew rate of this first power signal is intentionally limited to control EMI emissions. Each half circuit also has a transient power supply which may be selectively engaged to provide a second power signal to the amplifier when the first power signal is insufficient to power the amplifier. Each half circuit may also include a low voltage power supply which provides a third power signal to the amplifier, allowing the main power supply to be disabled when a low power level is required, further reducing EMI emissions. Each half circuit has a control circuit which regulates the power output from the main and transient power supplies. The control circuit may provide a pulse width modulated control signal or a pulse density modulated control signal to control the switching regulator. If a pulse density control signal is provided, the switching regulator may be a resonant switching regulator. The power amplifier may be modified for use with a bridge amplifier, with multiple channels and may incorporate an overload detection circuit. In another embodiment of power amplifier the transient power supply is replaced with a transient control circuit that, when the first power supply is insufficient to power the amplifier, temporarily forces the first power supply to a 100% duty cycle, and then for a longer period, increases the duty cycle from its normal level to allow the first power supply to adequately power the amplifier more quickly.

FIELD OF THE INVENTION

[0001] This invention relates to power amplifiers. More particularly, the invention relates to power amplifiers which efficiently amplify one or more input signals with a large dynamic range while producing low electromagnetic emissions.

BACKGROUND OF THE INVENTION

[0002] A music audio signal or a movie soundtrack typically has a large dynamic range. Such signals often have a peak-to-average magnitude ratio of 8-to-1 or even higher. In addition, peaks in such signals are relatively infrequent and at most times, the signal has a magnitude close to its average magnitude. A power amplifier for such a signal must be capable of producing an output signal with corresponding high peaks and with a comparatively low average magnitude. A number of power amplifiers are known which vary the power supplied to the power amplifier's main amplification circuit (or amplifier) so that there is limited headroom between the supplied power and the magnitude of the power amplifier's output signal.

[0003] For example, U.S. Pat. No. 3,772,606 describes a linear class H power amplifier with four fixed power rails. Two of the rails are used to supply power to the amplifier of the power amplifier during positive half waves of the output signal and the other two rails are used to supply power to the power amplifier during negative half waves of the output signal. Of the two rails used to supply the main amplifier during the positive half waves of the output signal, one is a low voltage rail and the other is a high voltage rail. Only the low voltage rail is used to power the amplifier when the output signal has a magnitude well below that of the low voltage rail. As the output signal approaches the low voltage rail, the high voltage rail is turned on to supply additional power. This device effectively reduces the average voltage drop (i.e. the headroom) from the supply rail to the output signal, thereby improving the efficiency of the power amplifier. However, this solution is far from ideal, especially where the output signal has an average level substantially less than the lower supply rail, or slightly higher than the lower supply rail. In either case, there will still be substantial headroom between the output signal and the power supplied to the amplifier.

[0004] U.S. Pat. No. 4,430,625 describes a power amplifier that addresses this problem by providing low voltage rails which have a variable magnitude proportional to the magnitude of the output signal. The low voltage rails are provided by a switching regulator and their magnitude is controlled using a fixed frequency pulse width modulated (PWM) control signal. When the low voltage rails are insufficient to power the amplifier, high voltage rails provided by fast acting linear regulators are utilized to make up for the deficiency. This device further reduces the headroom between the output signal and the power supply to the amplifier when the output signal is lower than the low voltage rails. However, it is susceptible to high electromagnetic interference (EMI) emissions due to its hard-switching low voltage regulators. In addition, this device has no mechanism for predicting the power required by the amplifier to generate the output signal at any particular time. This results in the power supply from the switching regulators being deficient when the input signal rises rapidly and in the worst case may cause the switching regulators to be deficient during every half wave (or during many half waves) of the output signal. This in turn leads to overuse of the linear regulators, increasing the power consumption of the power amplifier and decreasing its overall efficiency.

[0005] U.S. Pat. No. 5,347,230 describes a power amplifier which attempts to reduce the usage of the linear regulators by monitoring the current in the linear regulator and controlling the output of the switching regulator in a way that minimizes the current drawn from the linear regulator. The control circuit of this power amplifier is responsive to changes in the output signal to vary the power provided by the switching regulators. This design, which is responsive rather than predictive, leads to a slow response time for the switching regulators, possibly resulting in increased usage of the linear regulators. Furthermore, this device utilizes fast-switching switching regulators which generate large EMI emissions. In addition, this device suffers from a load dumping problem which may force a high current from a current source through a high impedance load, resulting in a large voltage spike across the load.

[0006] None of these devices is well suited for use with multiple channels. Most modern audio amplifiers produce at least five output channels (i.e. surround sound systems) and many produce six or more output channels (including a sub-woofer output). This is in contrast to the two channel systems (i.e. left and right signals) which were common in the past. Providing five or more duplicate power supply circuits for each power amplifier within a single audio amplifier increases both the size and cost of the audio amplifier.

[0007] Furthermore, none of these devices provide for protection of the amplification from over-current, over-temperature or other overload conditions. Such protection is essential for practical commercial use of a power amplifier circuit.

[0008] Accordingly, there is a need for a power amplifier for audio signals that provides an efficient power supply with low EMI emissions and with low headroom between the power supplied to the amplification circuit and the output signal of the power amplifier. It is preferable if the power amplifier has a predictive control system that allows the headroom to be reduced while ensuring that sufficient power is provided to the amplification circuit (or circuits) at all times. It is also preferable that the control circuit and regulation system of the power amplifier be adaptable for use with multiple channels. It is also desirable that the power amplifier be adaptable to protect the amplification circuit of each channel so as to prevent the amplification circuit from being damaged by over-current, over-temperature or other overload conditions.

SUMMARY OF THE INVENTION

[0009] In a first embodiment, present invention provides: a power amplifier for receiving an input signal at an input terminal and producing an output signal at an output terminal, said output signal corresponding to said input terminal, said power amplifier having a first power supply circuit comprising: an amplifier coupled to said input terminal for receiving said input signal and coupled to said output terminal for providing said output signal, said amplifier having a power input terminal for receiving a power input signal; a switching regulator coupled to said power input terminal for providing a switching power signal to said amplifier, wherein said switching power signal forms a first part of power input signal; a linear regulator coupled to said power input terminal, said linear regulator being selectively engageable to provide a linear power signal to said amplifier, wherein said linear power signal forms a second part of said power input signal; an input signal processing circuit coupled to said input terminal for receiving said input signal and for providing a rectified signal indicating the amount of power required by said amplifier; a control circuit coupled to said input signal processing circuit and to said power input terminal for controlling said switching power signal and said linear power signal in response to an error signal corresponding to a between said rectified signal and said power input signal; a linear regulator control circuit coupled to said input signal processing circuit for receiving said rectified signal and coupled to said linear regulator for controlling the engagement of said linear regulator in response to said rectified signal.

[0010] In a second embodiment, the present invention provides a power amplifier for receiving an input signal at an input terminal and producing an output signal at an output terminal, said output signal corresponding to said input terminal, said power amplifier having a first power supply circuit comprising: an EMI isolation circuit coupled to said input terminal for receiving said input signal and to an internal input node for providing an EMI-decoupled signal corresponding to said input signal; an amplifier coupled to said input terminal for receiving said input signal and coupled to said output terminal for providing said output signal, said amplifier having a power input terminal for receiving a power input signal; a switching regulator coupled to said power input terminal for providing a switching power signal to said amplifier, wherein said switching power signal forms a first part of power input signal; a linear regulator coupled to said power input terminal, said linear regulator being selectively engageable to provide a linear power signal to said amplifier, wherein said linear power signal forms a second part of said power input signal; an input signal processing circuit coupled to said internal input node for receiving said EMI-decoupled signal and for providing a rectified signal indicating the amount of power required by said amplifier; a control circuit coupled to said internal input signal processing circuit and to said power input terminal for controlling said switching power signal and said linear power signal in response to an error signal corresponding to a between said rectified signal and said power input signal; a linear regulator control circuit coupled to said input signal processing circuit for receiving said rectified signal and coupled to said linear regulator for controlling the engagement of said linear regulator in response to said rectified signal.

BRIEF DESCRIPTION OF THE DRAWINGS

[0011] The present invention will be described, by way of example only, with reference to the drawings, in which:

[0012]FIG. 1 is a block diagram of a first embodiment of a power amplifier according to the present invention;

[0013]FIG. 2 illustrates a second embodiment of a power amplifier according to the present invention;

[0014]FIG. 3 illustrates a third embodiment of a power amplifier according to the present invention;

[0015]FIG. 4 illustrates a fourth embodiment of a power amplifier according to the present invention;

[0016]FIG. 5 illustrates a illustrates the headroom between a power supply signal of the power amplifier of FIG. 4 and a pair of input signals to that power amplifier;

[0017]FIG. 6 illustrates a fifth embodiment of a power amplifier according to the present invention;

[0018]FIG. 7 illustrates the relationship between power signal produced by a switching regulator and a low voltage power supply of the power amplifier of FIG. 6;

[0019]FIG. 8 illustrates a sixth embodiment of a power amplifier according to the present invention;

[0020]FIG. 9 illustrates a pulse of a power signal produced by a resonant switching regulator of the power amplifier of FIG. 8;

[0021]FIG. 10 illustrates a seventh embodiment of a power amplifier according to the present invention;

[0022]FIG. 11 illustrates part of the feedback control circuit of the power amplifier of FIG. 10;

[0023]FIG. 12 illustrates another part of the feedback control circuit of the power amplifier of FIG. 10;

[0024]FIG. 13 is a timing diagram illustrating the production of PDM control signal of the power amplifier of FIG. 10;

[0025]FIG. 14 illustrates an eighth embodiment of a power amplifier according to the present invention;

[0026]FIG. 15 illustrates an input filtration circuit according to the present invention which may be used with a power amplifier;

[0027]FIG. 16 is a block diagram of a ninth embodiment of a power amplifier according to the present invention;

[0028]FIG. 17 illustrates a tenth embodiment of a power amplifier according to the present invention; and

[0029]FIG. 18 is a timing diagram illustrating the operation of the power amplifier of FIG. 17.

DETAILED DESCRIPTION OF EXEMPLARY EMBODIMENTS

[0030] Several exemplary power amplifiers made according to the present invention will now be described. Corresponding components of each power amplifier are identified by the same or similar reference numerals.

[0031] Reference is first made to FIG. 1, which is a block diagram of a power amplifier 100 according to the present invention. Power amplifier 100 has an input terminal 102, a positive half circuit 108, an amplifier 104, and output terminal 106 and a negative half circuit 110.

[0032] Input terminal 102 is configured to receive an input signal 130. Amplifier 104 is coupled to input terminal 102 to receive input signal 130 and to provide a corresponding output signal 132 at an output terminal 106. A load 134 is coupled to output terminal 106 to receive output signal 132.

[0033] Positive half circuit 108 has an input signal compensation block 105, a summer 137, a control circuit 116, a main power supply 118, a transient detect block 119, a transient power supply 123 and an output power signal compensation block 135. Negative half circuit 110 has the same structure as positive half circuit 108 and has a complementary operation. Only positive half circuit 108 will be described in detail.

[0034] Input compensation block 105 is coupled to input terminal 102 to receive input signal 130 and to provide a compensated input signal 140 at a terminal 117. Amplifier 104 receives a power signal V_(t) from a positive power terminal 136. Power signal V_(t) is the sum of a power signal V_(s) provided by main power supply 118 and a power signal V_(l) provided by transient power supply 123. The voltage of power signal V_(t) is equal to the greater of the voltages of power signals V_(s) and V_(l). Output power signal compensation block 135 is coupled to positive power terminal 136 and provides a compensated power signal V_(tr), which has a range comparable to that of compensated input signal 140. Summer 137 compares compensated input signal 140 to compensated power signal V_(tr) and provides an error signal 139. Input compensation block 105 and output power signal compensation block 135 are configured to produce.

[0035] Control circuit 116 receives error signal 139 and produces a first control signal 142 to control main power supply 118. Main power supply 118 is responsive to first control signal 142 and produces power signal V_(s) with a magnitude corresponding to control signal 142.

[0036] At any particular time when power amplifier 100 is in operation, amplifier 104 will require sufficient power V_(req) to produce output power signal 132. The specific amount of power V_(req) required will vary depending on the magnitude of input signal 130, the amount of amplification desired and power required to power the components of amplifier 104. Any excess power supplied to amplifier 104 will be dissipated in amplifier 104. Such dissipated power is lost and increases the power consumption of power amplifier 100. Accordingly, it is desirable to reduce the headroom between power signal V_(t) and the required power level V_(req). For reasons explained below, it is desirable to provide a safety margin between V_(t) and V_(req).

[0037] Input compensation block 105 and output power signal compensation block 135 are configured to produce compensated input signal 140 and compensated power signal V_(tr) such that error signal 139 will be zero or almost zero (i.e. compensated input signal 140 will be approximately equal to compensated power signal V_(tr)) when output power signal V_(t) has a magnitude V_(targ) that is slightly greater than V_(req). Control circuit 116 produces control signal 142 so that main power supply 118 produces power signal V_(s) with a magnitude equal to V_(targ). This condition is the “normal operation” of power amplifier 100. Under normal operation, transient power supply 123 is inoperative.

[0038] Power signal V_(s) thus follows the waveform of output signal 132 with a headroom of V_(targ)−V_(req), during the normal operation of power amplifier 100. For various reasons (described below), it may be desirable to reduce the stewing rate of main power supply 118. This can have the result that, if a large transient occurs in the input signal 130, power signal V_(s) will not be able to track output signal 132. Power signal V_(s) may have a magnitude less that V_(targ) or even less than V_(req) for a finite time.

[0039] During this finite time, power amplifier 100 enters a “transient operation” and transient power supply 123 is engaged to provide power signal V_(l) to amplifier 104. Transient detection block 119 monitors compensated input power signal 140 at terminal 117 and, when a transient that exceeds a selected threshold occurs in input signal 130, transient detection block 119 enables transient power supply 123. The threshold is selected to that transient power supply 123 will be engaged when main power supply 118 is unlikely to be able to provide power signal V_(s) with a magnitude approximately equal to V_(targ). The selection of the threshold will depend on the slew rate of main power supply 118.

[0040] Transient power supply 123 has a very short slewing time and produces power signal V_(l) in response to a second control signal 144 provided by control circuit 116. Control circuit 116 configures control signal 144 so that power signal V_(l) will be approximately equal to V_(targ). (The magnitude of power signal V_(l) is discussed in detail below.) When transient detection block 119 enables transient power supply 123, transient power supply 123 quickly produces power signal V_(l) approximately equal to V_(targ). Power signal V_(t) thereby has a magnitude approximately equal to V_(targ), providing approximately the desired amount of headroom for amplifier 104.

[0041] Reference is next made to FIG. 2, which is a block diagram of a second power amplifier 200 made according to the present invention. Power amplifier 200 has an input terminal 202, an amplifier 204, an output terminal 206, a positive half circuit 208 and a negative half circuit 210.

[0042] Input terminal 202 is configured to receive an input signal 230. Amplifier 204 is coupled to input terminal 202 to receive input signal 230 and to provide an amplified output signal 232 corresponding to input signal 230 at output terminal 206. A load 234 is coupled to output terminal 206 to receive output signal 232. Amplifier 204 receives power from a positive power input terminal 236.

[0043] Only the positive half circuit 208 of power amplifier 200 will be described in detail here. Positive half circuit 208 is operative to power amplifier 204 during positive half waves of output signal 232. Negative half circuit 210 has the same construction as positive half circuit 208 and has a complementary operation, providing power to amplifier 204 during negative half waves of output signal 232.

[0044] Positive half circuit 208 includes an input signal compensation block 205, a transient detect block 219, a control circuit 216, a power source 212, a main power supply 218, a transient power supply 223, a summer 237, and an output power signal compensation block 235. Input signal compensation block 105 includes an offset block 211, frequency compensation block 214 and a rectifier 215. Transient detect block includes a peak detector 220, a differentiator 222 and a comparator 255. Main power supply 218 is a switching regulator. Transient power supply 223 includes a switch 224 and a linear regulator 226. Output power signal compensation block 235 comprises an amplifier.

[0045] Offset block 211 is coupled to input terminal 202 to receive input signal 230. Offset block 211 adds a relatively small selected offset to input signal 230 to provide an offset input signal 231. Typically, the offset added by offset block 211 will be less than 10% of the expected 0-to-peak range of input signal 230 and is selected to ensure that the power provided to amplifier 204 by positive half circuit 208 is slightly higher than is actually required by amplifier 204 to generate output signal 232 (i.e. to ensure that V_(targ) is slightly higher than V_(req)).

[0046] Frequency compensation block 214 is coupled to offset block 211 to receive offset input signal 231. Frequency compensation block 214 provides a frequency compensated signal 238 corresponding to offset input signal 231 to rectifier 215. Frequency compensation block 214 may be implemented as a phase lead network to (i) increase the amplitude of frequency compensated signal 238 at higher frequencies and to (ii) phase advance the voltage of frequency compensated signal 238 with respect to its current, as compared to the phase angle of input signal 230.

[0047] Rectifier 215 receives frequency compensated signal 238 and provides a corresponding half wave rectified signal 240 at terminal 217. Rectifier 215 is a half wave rectifier which essentially discards negative portions of frequency compensated signal 238. The negative portions frequency compensated signal 238 are not required in positive half circuit 208 since positive half circuit 208 provides power to amplifier 204 only during positive half waves of output signal 232.

[0048] Rectified signal 240, which is analogous to the compensated input signal 140 of power amplifier 100 (FIG. 1), corresponds to the magnitude of offset input signal 231, and therefore corresponds generally to the magnitude of output signal 232 and to the power required by amplifier 240 to generate output signal 232 during a positive half wave of input signal 230. The offset added to offset input signal 231 by offset block 211 results in rectified signal 240 reflecting a slightly higher power level than is actually required by amplifier 204.

[0049] Amplifier 204 receives a power signal V_(t) from a positive amplifier power terminal 236. Power signal V_(t) is the sum of a power signal V_(s) provided by main power supply 218 and a power signal V_(l) provided by transient power supply 223. Output power signal compensation block 235 is coupled to terminal 236 to receive power signal V_(t) and provide a reduced power signal V_(tr), which has a range comparable to rectified signal 240. Reduced power signal V_(tr) has the same magnitude as rectified signal 240 when power signal V_(t) is providing sufficient power for amplifier 204 to generate the required output signal 232 with approximately the headroom configured by offset block 211.

[0050] Summer 237 receives rectified signal 240 and subtracts reduced power signal V_(tr) to provide an error signal 239. At any point in time, error signal 239 represents the difference between the power required by amplifier 204 to produce output signal 232 (i.e. as represented by rectified signal 240) and the power presently being supplied to amplifier 204 (i.e. power signal V_(t)).

[0051] Control circuit 216 is coupled to summer 237 to receive error signal 239. Control circuit 216 provides a first control signal 242 at a first control terminal 241 and a second control signal 244 at a second control terminal 243 in response to error signal 239. In power amplifier 200, control signal 242 is a pulse width modulated (PWM) control signal.

[0052] Main power supply 218 is coupled to power source 212 to receive electrical power. Power source 212 has an output voltage of V_(max). Main power supply 218 is a switching regulator and includes a switch 246 and a diode 248. Switch 246 is coupled to control terminal 241 and is responsive to PWM control signal 242 to produce a power signal V_(si). Diode 248 ensures that current may flow in main power supply 218 at all times (i.e. when switch 246 is open, current may flow from ground through diode 248). Main power supply 218 also includes an integrating LC filter 250, which smooths power signal V_(si) to produce power signal V_(s), which is delivered to positive power terminal 236 through diode 251.

[0053] Amplifier 204 receives power signal V_(s) from positive power terminal 236. Amplifier 204 requires sufficient power V_(req) to produce output signal 232 and to power the components of amplifier 204 itself. Due to the small offset added by offset block 211, error signal 239 indicates that a slightly larger power signal is required by amplifier 204 than the power V_(req) that is actually required. As a result, control circuit 216 sets control signal 242 such that main power supply 218 will produce a power signal V_(targ), which is slightly higher than the power V_(req) required by amplifier 204. Amplifier 204 will therefore ideally have a headroom equal to V_(targ)−V_(req).

[0054] Due to the phase lead effect of frequency compensation block 214, rectified signal 240 and control signal 242 will be advanced compared to the time at which amplifier 204 will actually receive the corresponding part (i.e. the current) of input signal 230. As a result, main power supply 218 will begin to shift its output power signal V_(s) to the power level V_(req) required by amplifier 204 before amplifier 204 actually requires that level of power. In this way, the phase lead effect of frequency compensation block 214 gives main power supply 218 more time to respond to changes in the power level required by amplifier 204 to produce output signal 232. A similar predictive effect could be achieved by removing frequency compensation block 238 and differentiating rectified signal 240 with respect to time before it is fed to summer 239. This approach would result in a change in the waveform of the output power signal V_(s) of main power supply 218 since rectified signal 240 is not a sinusoidal signal.

[0055] The small excess power (i.e. the headroom) in power signal V_(s) (V_(targ)−V_(req)) is dissipated in amplifier 204. Ideally, the excess power in power signal Vs would be zero or almost zero, although such an ideal result cannot practically be achieved. One reason for this is that power signal V_(s) will exhibit some degree of ripple. Power signal V_(si) is produced by the operation of switch 246 and will accordingly have a fairly high degree of ripple and may actually have substantial high frequency components. Filter 250 operates to integrate power signal V_(si) and smooth it thereby producing power signal V_(s). However, power signal V_(s) will still exhibit some ripple, and it is necessary to ensure that even when the ripple causes power signal V_(s) to be below V_(targ), it is still sufficient to power amplifier 204. For example, if output signal 232 has a magnitude of 10 volts at a particular point in time and exhibits a symmetrical ripple of 5 volts, then V_(targ) must be set to at least 12.5 volts so that even when power signal V_(s) is at it lowest level (during that particular time), which will be 12.5-2.5 volts, it will provide sufficient power to amplifier 204. More practically, in this example, V_(targ) may be set to 13.5 volts or more to provide an additional safety margin, to accommodate for the voltage drop across diode 251 and to provide power for the components of amplifier 204.

[0056] The combination of the effect of offset block 211 and the phase lead effect of frequency compensation block 214 is to configure main power supply 218 to produce a power signal V_(s) which has a slightly higher magnitude than is required by amplifier 204, and which is generally matched in time to the power needs of amplifier 204. A person skilled in the art will be capable of adjusting the offset added by offset block 211 and the magnitude of the phase lead effect of frequency compensation block 214 to provide an appropriate power signal V_(s).

[0057] When the frequency of input signal 230 is relatively low, power signal V_(s) will follow the waveform of output signal 232, with some headroom. As the slew rate of input signal 230 rises, main power supply 218 may be unable to change the magnitude of power signal V_(s) sufficiently quickly to follow output signal 232, which will have a correspondingly high slew rate. (Main power supply 218, which is a switching power regulator, has an intentionally limited slew rate, which is discussed below, increasing the possibility that this will occur.) When this occurs, power signal V_(s) will follow the envelope of output signal 232. However, due to the slow slew rate of main power supply 218 and the high slew rate of the output signal 232, output signal V_(s) may not have any headroom between it and the envelope of output signal 232. In fact, output signal V_(s) may be able to follow only the average of the envelope of output signal 232 and may actually be lower than output signal 232 at times, and may therefore be insufficient to power amplifier 204.

[0058] As mentioned above, frequency compensation block 214 increases the amplitude of frequency compensated signal 238 at higher frequencies. As a result, when higher frequencies are present in input signal, rectified signal 240 will be magnified in comparison to input signal 230, and therefore, control circuit 216 will regulate main power supply 218 to a higher target power level V_(targ). If the increased amplitude of frequency compensated signal 238 at high frequencies is sufficient, then main power supply 218 may provide a sufficient power signal V_(s) to power amplifier 204.

[0059] Another mechanism for ensuring that main power supply 218 produces a power signal V_(s) that is sufficient to power amplifier 204 may be incorporated into offset block 211. In power amplifier 200, offset block 211 adds a fixed offset to input signal 230 to produce offset fixed signal 231. In an alternative embodiment of a power amplifier according to the present invention, offset block 211 may be configured to detect the frequency components on input signal 230 and to add a smaller offset to input signal 230 if the highest frequency component is lower than a selected frequency and to add a larger offset to input signal 230 if frequency components exceeding equal to or exceeding the selected frequency exist. This will result in a larger power signal V_(s) when input signal 230 has frequency components higher than the selected frequency, thereby providing a higher headroom between the required power level V_(req) and the target power level V_(targ).

[0060] One skilled in the art will recognize that in power amplifier 200, the addition of a fixed offset to input signal 230 will cause a part of each negative half wave of input signal 230 (immediately before the beginning and after the end of each positive half wave) to have a magnitude greater than 0, which then results in that part of each negative half wave being treated as part of a positive half wave of input signal 230. Conversely, the complementary operation of negative half circuit 210 will cause similar parts of the positive half wave of input signal to be treated as part of the negative half waves. This may be acceptable and even preferable depending on the desired performance of a power amplifier according to the present invention. If this is not desired, offset block 211 may be coupled to the output of rectifier 215 to allow rectifier 215 to remove all parts of each negative half wave. Offset block 211 will then add the fixed offset to rectified signal 240.

[0061] Returning to a description of power amplifier 200, main power supply 218 will emit some electromagnetic radiation, which produces electromagnetic interference (EMI) in nearby electronic devices. As is well understood, the amount of EMI produced by a switching regulator such as main power supply 218 depends on its switching rate. In order to reduce EMI, the switching rate of switch 246 is reduced by selecting a relatively low frequency for PWM control signal 242. The precise frequency chosen for PWM control signal (which is a fixed frequency signal) will depend on the characteristics of load 234, output signal 232 and energy which must be delivered to properly power amplifier 204, among other criteria. A person skilled in the art will be capable of selecting a suitable frequency to balance these considerations with the need to reduce EMI emissions in a particular implementation of the present invention. Reducing the frequency of PWM control signal 242 and the switching rate of switch 246 has several effects.

[0062] First, it increases the ripple in power signal V_(si). To counteract this effect, the time constant of LC filter 250 is increased so that power signal V_(s) will be effectively smoothed despite the relatively low frequency at which switch 246 is operated.

[0063] Second, the rate at which the magnitude of power signal V_(s) can rise or fall (i.e. the slew rate of power signal V_(s)) is reduced. If input signal 230 has a rapid increase in magnitude, amplifier 204 will require a rapid increase in its power supply to properly generate output signal 232 (i.e. V_(targ) will rise rapidly). Due to its slow slew rate, main power supply 218 may be unable to increase the magnitude of power signal V_(s) sufficiently quickly to power amplifier 236. Eventually, power signal V_(s) will rise to V_(targ), up to a maximum of V_(max) volts. However, this will take a finite time.

[0064] During this finite time period, additional power is provided by transient power supply 223 to amplifier 204 as follows. Transient detect block 219 operates to provide a digital transient signal 257. Peak detector 220 is coupled to rectifier 215 to receive rectified signal 240. Peak detector 220 provides a peak signal 252 corresponding to the peak envelope of the rectified signal 240 to differentiator 222. Differentiator 222 differentiates peak signal 252 to provide a differentiated signal 254. The magnitude of differentiated signal 254 at any point in time is indicative of the rate at which input signal 230 is changing. Comparator 255 compares differentiated signal 254 with a threshold voltage 256 to provide transient signal 257. When differentiated signal 254 exceeds threshold 256, transient signal 257 is high; otherwise, it is low.

[0065] Switch 224 is responsive to transient signal 257 to selectively connect control circuit 216 to linear regulator 226. Linear regulator 226 may be a power MOSFET. The drain of linear regulator 226 is connected to power source 212 and the source of linear regulator 226 is connected to positive power input terminal 236. The gate of linear regulator 226 is connected to a terminal a of switch 224. Terminal b of switch 224 is coupled to control circuit 216 and terminal c of switch 224 is coupled to ground.

[0066] When transient signal 257 is high, switch 224 engages transient power supply 224 by coupling the gate of linear regulator 226 to control terminal 243 (i.e. terminal a is coupled to terminal b). At all other times, switch 224 couples the gate of linear regulator 226 to ground (i.e. terminal a is coupled to terminal c).

[0067] Like control signal 242, control signal 244 is set by control circuit 216 so that transient power supply 223 will produce power signal V_(l) equal to V_(targ). (It may be desirable to make power signal V_(l) slightly lower than power signal V_(s). This is discussed further below.) Also, control signal 244 will be advanced compared to the time at which amplifier 204 will actually require the power level indicated by control signal 244. Control signal 244 is an analog signal, as opposed to control signal 242, which is a PWM signal.

[0068] When the gate of linear regulator 226 is coupled to control terminal 243, it receives control signal 244. Linear regulator 226 is responsive to control signal 244 to provide power signal V_(l) to positive power input terminal 236. Linear regulator 226 has a very fast slew rate, and therefore its power signal V_(l) rises quickly to V_(targ).

[0069] Eventually, the magnitude of power signal V_(s) will rise to V_(targ). At this point, power signal V_(s) will be sufficient to separately power amplifier 204. Until this time, it is necessary for switch 224 to keep transient power supply 223 engaged. The period for which switch 224 must remain in this state will depend on the slew rate of main power supply 218 and on the rate of change of input signal 230. The slew rate of main power supply 218 may be calculated (or more likely, the slew rate may be pre-determined based on the maximum EMI that power amplifier 200 is permitted to emit and then main power supply 218 may be designed to have the selected slew rate). Threshold 256 is selected to correspond to this slew rate, so that main power supply 223 will be kept engaged by switch 224 until power signal V_(s) is able to rise to V_(targ).

[0070] Once transient power supply 223 is disengaged, it is important that peak detector 220 and differentiator 222 operate properly to ensure that linear regulator 224 may be engaged again if input signal 230 has another rapid rise in magnitude. This may be done by setting the discharge rate of peak detector 220 to correspond to the slew rate of main power supply 218 (which will depend on time constant of filter 250 and will also be affected by delays through the feedback network comprising amplifier 235, summer 237 and control circuit 216). This will ensure that the magnitude of peak signal 252 will fall as fast as power signal V_(s), and will ensure that differential signal 254 will exceed threshold 256 when input signal 230 rises too quickly for main power supply 218 to supply enough power to amplifier 204. Furthermore, this will also ensure that differential signal 254 will not fall below threshold 256 too quickly, causing linear regulator 226 to be disengaged before power signal V_(s) has risen to V_(targ).

[0071] As noted above, control signals 242 and 244 are set so that main power supply 218 and linear regulator 226 will provide power signals V_(s) and V_(l) of approximately the same magnitude (V_(targ)). This ensures that power signal V_(t) will have a smooth transition when linear regulator 226 is engaged by the operation of switch 224. When main power supply 218 is unable to raise its output power signal V_(s) sufficiently to power amplifier 204, differentiated signal 254 will exceed threshold 256. Very quickly thereafter, switch 224 will engage transient power supply 223. Since transient power supply 223 is configured to produce the same power signal V_(targ) as main power supply 218, and since linear regulator 226 is engaged very quickly after main power supply 218 becomes unable to produce the configure power signal V_(targ), engaging transient power supply 223 will not cause a substantial jump in the total power signal V_(t) supplied to amplifier 204. Similarly, when transient power supply 223 is disengaged (when differentiated signal 254 falls below threshold 256), both transient power supply 223 and main power supply 218 will be generating the same power signal V_(targ) (i.e. V_(s)=V_(l)=V_(targ)). As a result, there will be a smooth transition in V_(t) when transient power supply 223 is switched off.

[0072] Control signal 244 cannot be configured to produce a power signal V_(l) with a larger magnitude than power signal V_(s) (i.e. V_(l) cannot be larger than V_(targ)). If this occurs, power signal V_(s) produced by main power supply 218 may not reach V_(targ) before transient power supply 223 is disengaged, and therefore, may not be able to power amplifier 204 at that time. If power signal V_(l) exceeds the level to which main power supply 218 is being regulated by control signal 242 (i.e. V_(l) exceeds V_(targ)), then the magnitude of power signal V_(t) will be equal to the magnitude of power signal V_(l). Since V_(t) exceeds V_(targ), error signal 239 will indicate that V_(targ) may be lowered. Control circuit 216 will accordingly set control signal 242 to reduce power signal V_(s). Accordingly, control circuit 216 must ensure that control signal 242 remains constant when transient power supply 223 is engaged. This may be done by setting control signal 244 to operate linear regulator 226 to produce power signal V_(l) with a slightly smaller magnitude than power signal V_(s) produced by main power supply 218 under the control of control signal 242 (i.e. V_(l) may be slightly lower than V_(targ)). Although this will cause a small transition in power signal V_(t) whenever transient power supply 223 is engaged, it will not have a substantial effect when transient power supply 223 is disengaged.

[0073] Power amplifier 200 provides an efficient circuit for receiving input signal 230 to provide an amplified output signal 232. Amplifier 204, which performs the amplification of input signal 230, receives power from main power supply 218 and from linear regulator 226. At most times, amplifier 204 receives power only from main power supply 218 and transient power supply 223 is disengaged by grounding its control terminal (i.e. its gate). The power output signal V_(s) of main power supply 218 is configured to follow and slightly exceed the power V_(req) required by amplifier 204 to generate output signal 232. This reduces power dissipation in amplifier 204 and the overall power consumption of power amplifier 200. The switching frequency of main power supply 218 is kept intentionally low to reduce EMI generation. This causes main power supply 218 to have a slow slew rate and main power supply 218 may not be able to provide sufficient power to amplifier 204 to produce output signal 232. When such a condition occurs, transient power supply 223 is engaged. Transient power supply 223 has a fast slew rate, allowing it to quickly provide the power V_(req) required by amplifier 204 to produce output signal 232. Transient power supply 223 is kept engaged for a period sufficient to allow main power supply 218 to increase its power supply signal V_(s) to V_(req), and then transient power supply 223 is disengaged.

[0074] Reference is next made to FIG. 3, which is a block diagram of a third power amplifier 280 according to the present invention. Power amplifier 280 is similar to power amplifier 200 except that the control signals for main power supply 218 and transient power supply 223 are generated by separate control circuits. In addition to the components of power amplifier 200, power amplifier 280 has a first control circuit 282, a second control circuit 283, an amplifier 294 and a summer 286.

[0075] Amplifier 294 is coupled to main power supply 218 to receive power signal V_(s). Amplifier 294 reduces the range of power signal V_(s) and produces a reduced power signal V_(sr), which has a range comparable to that of rectified signal 240. Summer 286 is coupled to rectifier 215 and to amplifier 294 to receive rectified signal 240 and reduced power signal V_(sr). Summer 286 produces an error signal 288 which indicates the difference between the power V_(req) required to produce output signal 232 (i.e. as represented by rectified signal 240) and the power signal V_(s) being supplied by main power supply 218 to amplifier 204. Control circuit 282 is coupled to summer 286 to receive error signal 288. Control circuit 282 provides a PWM control signal 290 in response to error signal 288 to control switch 246 of main power supply 218.

[0076] Amplifier 235 is coupled to terminal 236 to receive power signal V_(t) and operates as in power amplifier 200 to produce reduced power signal V_(tr). Summer 237 is coupled to rectifier 215 to receive rectified signal 240 and to amplifier 235 to receive reduced power signal V_(tr) and operates as in power amplifier 200 to produce error signal 239. Control circuit 284 receives control signal 239 and produces an analog control signal 292, which is similar to control signal 244 of power amplifier 200. Switch 224 may couple control signal 292 to the gate of linear regulator 226, when transient power supply 223 is required to supply power signal V_(l) to amplifier 204, in the same manner as described above in relation to power amplifier 200. Power signals V_(l) and V_(s) are diode or'd through diode 251 to produce power signal V_(t).

[0077] By generating control signals 290 and 292 separately through independent feedback paths and control circuits 282 and 284, respectively, main power supply 218 and transient power supply 223 may be independently controlled. In particular, by referencing control signal 290 to the output of main power supply 218 only, it is not necessary to ensure that power signal V_(l) does not exceed the voltage V_(targ) to which main power supply 218 is being regulated, as is required in the case of power amplifier 200.

[0078] Reference is next made to FIG. 4, which illustrates a fourth power amplifier 300 according to the present invention. Power amplifier 300 illustrates the use of a power amplifier based on power amplifier 200 to power several channels simultaneously. Power amplifier 300 also illustrates the use of overload detectors to protect the amplifiers of power amplifier 300.

[0079] Power amplifier 300 has two input terminals 302 a, 302 b, two amplifiers 304 a, 304 b, a positive half circuit 308 and a negative half circuit 310. Power amplifier 300 is configured to amplify input signals 330 a and 330 b, which are received at input terminals 302 a and 302 b, respectively. Input signals 330 a and 330 b are two separate input signals and may be two channels in an audio system. For example, input signals 330 a and 330 b may be the left and right channels in a standard audio system.

[0080] Amplifiers 304 a and 304 b are coupled to input terminals 302 a and 302 b to receive input signals 330 a and 330 b. Amplifier 304 a and 304 b provide amplified output signals 332 a and 332 b, which correspond to input signal 302 a and 302 b, to loads 334 a and 334 b.

[0081] Offset block 311 a, frequency compensation block 314 a and rectifier 315 a, which comprise a first input signal compensation block 305 a, are coupled to input terminal 302 a in the same manner as offset block 211, frequency compensation block 214 and rectifier 215 are coupled to input terminal 202 of power amplifier 200. Rectifier 315 a provides a rectified signal 340 a corresponding to input signal 330 a at terminal 341 a. Similarly, offset block 311 b, frequency compensation block 314 b and rectifier 315 b, which comprise a second input signal compensation block 305 b, are coupled to input terminal 302 b and rectifier 315 b provides a rectified signal 340 b corresponding to input signal 330 b at terminal 341 b. Terminals 341 a and 341 b are coupled to terminal 341 through diodes 360 a and 360 b. Rectified signals 340 a and 340 b are diode-or'd by diodes 360 a and 360 b, providing a rectified signal 340 at terminal 341, which corresponds generally to either the highest magnitude of signals 330 a and 330 b at any particular time. Offset blocks 311 and 311 b may be configured to compensate for the diode drops caused by diodes 360 a and 360 b in rectified signals 340 a and 340 b.

[0082] Rectified signal 340 generally corresponds to highest of the power levels required by either amplifier 304 a or 304 b to produce output signal 332 a or 332 b. Summer 368 receives rectified signal 340 and a combined overload detect signal 366, which is explained below and provides an adjusted rectified signal 370, which is used to produce power signal V_(t) at positive power input terminal 236 in the same manner as rectified signal 240 is used in power amplifier 200.

[0083] Amplifier 304 a is coupled to power input terminal 236 through an overload detect block 362 a. Amplifier 304 a receives power signal V_(t) and provides output signal 332 a. Similarly amplifier 304 b is coupled to power input terminal 236 through an overload detect block 362 b and provides output signal 332 b. Overload detect blocks 362 a and 362 b are configured to detect overload conditions in their associated amplifiers 304 a and 304 b. For example, overload detect blocks 362 a and 362 b may be configured to detect over-current, over-temperature or other overload conditions. If overload detect block 362 a detects such a condition in its associated amplifier 304 a, then overload detect block 362 a will produce an overload signal 366 a, which corresponds to the magnitude of the overload condition detected. Similarly, overload detect block 362 b will produce an overload signal 366 b if an overload is detected in amplifier 304 b. Overload signals 366 a and 366 b are diode or'd through diodes 364 a and 364 b to produce a combined overload signal 366, which is subtracted from rectified signal 340 to produce adjusted rectified signal 343. Adjusted rectified signal 343 is used to control the magnitude of power signal V_(t). In this way, overload detect blocks 362 a and 362 b operate to reduce power signal V_(t) to protect amplifier 304 a and 304 b when an overload occurs.

[0084] Since input signals 330 a and 330 b are independent of one another, they may have different magnitudes at any point in time. Accordingly, output signals 332 a and 332 b will have different amplitudes and amplifiers 304 a and 304 b will have different power requirements. Power signal V_(t) is large enough to power the amplifier 304 a or 304 b with the largest power requirement at any particular (assuming that no overload condition exists). As a result, one of the amplifiers will receive more power than it requires and will dissipate the excess power.

[0085] It has been found that the dissipation of this excess power in one of the amplifiers 304 a or 304 b does not substantially reduce the average efficiency of power amplifier 300. Reference is made to FIG. 5, which shows output signals 322 a and 322 b and power signal V_(t). In the portion of the signals shown, output signal 322 b is experiencing a peak at time t₁. Output signal 322 a is at a typical or average level. A typical audio signal, such as a movie soundtrack or music, may have a peak-to-average magnitude ratio of 8:1 or higher. At time t₁, output signal 322 a may have a current of 2 amps and a magnitude of 5 volts while output signal 322 b may have a current of 14 amps and a magnitude of 35 volts. (Output signal 322 b has 49 times more power than output signal 322 a). Typically, output signal V_(t) may have an average magnitude of 41 volts (which is selected to account for any ripple in output signal V_(t), to power the components of both amplifiers 204 a and 204 b and to provide reasonable headroom in the amplifier 204 a or 204 b with the higher power requirements) when amplifiers 304 a and 304 b are called on to produce these output signals 332 a and 322 b. Accordingly, output signal 322 a has a headroom of 36 volt and will dissipate 72 watts of power (i.e. 36 volts×2 amps). Output signal 322 b has a headroom of 6 volts and will dissipate 84 watts (i.e. 6 volts×14 amps). Amplifier 304 b will therefore actually exhibit a high power loss, even though amplifier 304 a has a much larger headroom. As a result, the large headroom created in amplifier 304 a due to the peak of output signal 304 b does not substantially increase the average power dissipated by power amplifier 300 in comparison to the power dissipated in amplifier 304 b at the same time. Furthermore, peaks in a typical musical selection or a movie soundtrack occur relatively infrequently (and often occur on several channels simultaneously) and since the average level of a typical selection is generally less than ⅛th the level of the peaks in the selection, sharing main power supply 218 and transient power supply 223 between more than one amplifier does not result in a substantial change in the overall efficiency of power amplifier 300.

[0086] Power amplifier 300 allows main power supply 218 and transient power amplifier 223 to be shared between two channels having separate inputs, amplifiers and outputs. The design of power amplifier 300 is not limited to two channels and may be used to power any number of channels. For example, power amplifier 300 could be adapted to power 6 or more channels simultaneously for use with a typical “surround sound” soundtrack for a movie, which may have 5 or more audio channels and 1 or more additional sub-woofer channels.

[0087] Reference is next made to FIG. 6, which illustrates a fifth embodiment of a power amplifier 400 according to the present invention. Power amplifier 400 is similar to power amplifier 200. Power amplifier 400 has an input terminal 202, an amplifier 204, an output terminal 206, a positive half circuit 408 and a negative half circuit 410. In addition to the structure defined above in relation to positive half circuit 208 of power amplifier 200, positive half circuit 408 includes a low voltage power supply 402 and a diode 404. Control circuit 216 of power amplifier 200 has been replaced with control circuit 416. Main power supply 218 of power amplifier 200 has been replaced with main power supply 418, which is a switching power regulator with an LC filter 450.

[0088] Reference is briefly made to FIG. 2. As noted above, the switching speed of main power supply 218 is kept intentionally low to reduce the amount of EMI generated by main power supply 218. To reduce the amount of ripple in power signal V_(s) (as compared to the relatively large ripple in power signal V_(si)), the time constant of filter 250 is increased. Although this provides a smoother power signal V_(s), it also further reduces the slew rate of main power supply 218, thereby preventing power signal V_(s) from closely following output signal 232.

[0089] Referring again to FIG. 6, it is therefore desirable to decrease the time constant of filter 450 to allow power signal V_(s) to follow the output signal 232 closely. This has the disadvantage that the ripple produced in power signal V_(si) may not be effectively removed from power signal V_(s). Depending on the time constant of filter 250, the ripple in power signal V_(s) may be quite large. The selection of an optimal time constant will depend on the nature of input signal 230, the desired efficiency of power amplifier 400 and the desired EMI emission level for power amplifier 400. The selection of a low time constant for filter 450 has three adverse consequences.

[0090] First, when the magnitude of power signal V_(s) is relatively low, the ripple in power signal V_(s) will produce a proportionally large amount of energy to be stored in (and later dissipated from) capacitor 451 of filter 450. The magnitude of the ripple in power signal V_(s) will be relatively constant regardless of the magnitude of power signal V_(s). The energy stored in capacitor 451 is equal to ½ CV_(s) ². When power signal V_(s) has a relatively low magnitude (which will be the case much of the time, since peaks in a typical music signal are relatively infrequent), the ripple in power signal V_(s) will be relatively large, and will therefore cause a proportionally larger excess amount of energy to be stored in capacitor 451 during each pulse created in power signal V_(si) by the closing and opening of switch 246. This excess energy is later dissipated and is essentially wasted. This increases the power consumption of power amplifier 400.

[0091] Second, the difference between the voltage to which power signal V_(s) is regulated (V_(targ)) and the voltage required by amplifier 204 to produce output signal 230 must be increased to ensure that power signal V_(s) is sufficient at all times to power amplifier 204. The increased average headroom between power signal V_(s) and the power V_(req) required by amplifier 204 results in increased power dissipation in the amplifier 204 and a higher overall power consumption for power amplifier 400. This may be done by increasing the offset added into input signal 230 by offset block 211 or by modifying control circuit 216.

[0092] Third, the ripple in power signal V_(s) may actually be coupled to output signal 232 through amplifier 204 and may be audible, depending on the frequency of the ripple (which in turn will depend on the frequency at which switch 246 is operated). To reduce the effect of this ripple on power signal V_(s) when the magnitude of power signal V_(s) is low, it may be desirable to eliminate the use of main power supply 418 when output signal 232 is relatively low.

[0093] This is accomplished by using low voltage power supply 402 when the magnitude of output signal 232 is relatively small. Low voltage power supply 402 is coupled to positive power supply terminal 236 and provides a fixed voltage DC power signal V_(LV) to amplifier 204 at all times. Power signal V_(LV) is diode-or'd with power signal V_(s) and power signal V_(l) through diodes 404 and 251. In power amplifier 400, at any particular time, power signal V_(t) delivered to positive power input terminal 236 is equal to the highest of power signals V_(l), V_(s) and V_(LV).

[0094] Control circuit 416 of power amplifier 400 may be configured to keep switch 246 open when the magnitude of output signal 232 is less than the magnitude of power signal V_(LV). This may be done by setting control signal 442 to 0 during these periods and it effectively eliminates the proportionally high energy loss which would otherwise occur when power signal V_(s) has a low magnitude. Furthermore, this eliminates the generation of EMI by switching regulator 232 during these periods when it is effectively disengaged. Control circuit 416 will be able to determine when to engage and disengage main power supply 418 based on error signal 239. Since power signal V_(t) will have a minimum magnitude of V_(LV), error signal 239 may be negative when input signal 230 is very small (and correspondingly amplifier 204 requires little power to generate output signal 230. When error signal 239 is negative (or below a selected threshold which may be built into control circuit 416), control circuit 416 will disengage main power supply 418.

[0095] Since power signal V_(LV) has a fixed magnitude (even during negative half waves of input signal 230), it becomes the minimum magnitude that power signal V_(t) can have. This does not substantially reduce the efficiency power amplifier 400 because the large ripple introduced into power signal V_(s) by decreasing the time constant of main power supply 218 essentially forces power signal V_(s) (and hence power signal V_(t)) to have a minimum average at least as large as one-half of the peak-to-peak magnitude of the ripple.

[0096] Reference is made to FIG. 7, which illustrates an average positive half wave of output signal 232 and power signal V_(s), V_(LV) and V_(t) during the corresponding period. The magnitude V_(LV) of low voltage power supply 402 is selected to be a fraction of the average magnitude V_(s-avg) of power supply V_(s) during periods when output signal 232 is at an average level (i.e. a period during which no peaks occur). (During such periods, transient power supply 223 will generally not be engaged.) During the period shown in FIG. 7, transient power supply 223 is disengaged. Between times t₂ and t₃ and between times t₄ and t₅, the magnitude of output signal 232 is less than the magnitude of power signal V_(LV) and power signal V_(t) is equal to power signal V_(LV). As noted above, control circuit 216 may be configured to disengage main power supply 418 is disengaged by setting control signal 442 to 0. In power amplifier 400, control circuit 416 is configured to disengage main power supply 418 when the magnitude of output signal 232 is less than a selected threshold V_(d). When the magnitude of output signal 232 approaches threshold V_(d), main power supply 418 is engaged by control circuit 416 to produce power signal V_(s). Threshold V_(d) must be selected to ensure that main power supply 418 is engaged whenever power signal V_(LV) would be insufficient to power amplifier 204 to produce output signal 232. In selecting threshold V_(d), the desired headroom (defined above as V_(targ)−V_(req)) must be taken into account. When main power supply 418 is engaged, power signal V_(t) will be equal to the higher of power signal V_(LV) and V_(s).

[0097] The use of low voltage power supply 402 in power amplifier 400 allows a filter 450 with a relatively short time constant to be used, allowing power signal V_(s) (and consequently power signal V_(t)) to more closely follow output signal 232 than is possible in power amplifier 200. Furthermore, the use of low voltage power supply 402 allows main power supply 418 to be disengaged during periods when it would have a large proportional power loss and would generate relatively high EMI emissions.

[0098] Reference is briefly made to FIG. 2. PWM control signal 242 consists of a series of pulses which may have differing pulse widths. The rising edges of successive pulses (assuming that a pulse is measured between a rising edge and a falling edge) are spaced apart by a fixed time period. Switch 246 is closed when PWM control signal 242 is high and switch 246 is open when PWM control signal 242 is low. The opening and closing operations of switch 246 produce an essentially square wave power signal V_(si). Transitions in PWM control signal 242 occur without regard to the current flowing through switch 246. As a result, switch 246 may be opened while substantial currents are flowing through it, creating EMI emissions and high switching stresses and switching losses on the components on main power supply 218. (These currents are drawn by amplifier 204 to produce output signal 232.) These inefficiencies may be alleviated by opening and closing switch 246 only when the current being drawn from main power supply 218 is zero or almost zero.

[0099] Reference is next made to FIG. 8 which is a block diagram of a sixth embodiment of a power amplifier 500 according to the present invention. Power amplifier 500 is similar to power amplifier 400, except for the configuration and structure of its control circuit 516 and main power supply 518. Power amplifier 500 has an input terminal 202 and an amplifier 204 which operate as in power amplifiers 200 and 400 to produce an output signal 232 at a terminal 206. Load 234, which may be a speaker, receives output signal 232. Power amplifier 500 also has a positive half circuit 508, which supplies power to amplifier 204 at a power terminal 236 during positive half waves of input signal 230, and a negative half circuit 510 which has the same structure as positive half circuit 508 and a complementary operation to supply power to amplifier 204 during negative half waves of input signal 230.

[0100] Positive half circuit 508 is similar to positive half circuit 408 of power amplifier 400, except that control circuit 408 has been replaced with a pulse density modulation (PDM) control circuit 508 and main power supply 418 has been replaced with a main power supply 518 which is a resonant switching power regulator. Main power supply 518 may be any type of resonant switching regulator such as a zero-current-switching (ZCS) converter, a zero-voltage-switching (ZVS) converter, a zero-voltage-switching quasi-resonant converter (ZVS-QRC), a zero-voltage-switching multi-resonant converter (ZVS-MRC), a constant-frequency, a zero-voltage-switching quasi-resonant converter (CF-ZVS-MRC). Such converters are described in U.S. Pat. No. 4,720,668, entitled “Zero Voltage Switching Quasi Resonant Converters” and in U.S. Pat. No. 5,479,337, entitled “Very Low Power Loss Amplifier for Analog Signals Utilizing Constant-Frequency Zero-Voltage Switching Multi-Resonant Converter”. Such regulators have the advantage of lower EMI emissions and lower switching losses than the non-resonant main power supplies 218, 318 and 418 described above in respect of power amplifiers 200, 280, 300 and 400.

[0101] Main power supply 518 is a zero-current-switching (ZCS) switching regulator. Main power supply 518 includes a switch 546, diodes 547, 548 and a LC resonant tank 570 comprised of an inductor 572 and a capacitor 574, and an LC filter 550.

[0102] PDM control circuit 516 receives error signal 239 and provides a PDM control signal 542 at terminal 241 and second control signal 244 at terminal 243, which is the same as second control signal 244 provided by control circuit 216 of power amplifier 200. However, PDM control signal 542 is different from control signal 242, which is a pulse width modulated signal. Instead, PDM control signal 542 consists of a series of identical “high” pulses—each high pulse has an identical pulse length or a “constant on-time” (and an identical magnitude). After each pulse (i.e. after its falling edge), another pulse will not begin, and PDM control signal 542 will remain low, for a selected “minimum off-time”. The pulses may be spaced apart by different periods, depending on the magnitude of the target power signal V_(targ) needed to power amplifier 204 indicated by rectified signal 240. When a higher magnitude power signal is required, the time period between pulses is reduced, and vice versa. As a result, PDM control signal 542 has a “variable off-time” between pulses.

[0103] Main power supply 518 receives PDM control signal 542 and produces a power signal V_(s-res) in response. Power signal V_(s-res) is analogous to power signal V_(s) produced by main power supply 218 of power amplifiers 200, 280, 300 and 400. Power signal V_(s-res) is diode-or'd with power signal V_(l) and V_(LV) to produce power signal V_(t), which is received by amplifier 204 at terminal 236.

[0104] Switch 546 is preferably a MOSFET type transistor which receives PDM control signal 542. Alternatively, another semiconductor device such as an IGBT, BJT or an SCR may be used. Switch 546 is open (i.e. the MOSFET is “off”) when PDM control signal 542 is low and is closed (i.e. the MOSFET is “on”) when PDM control signal 542 is high. As is known, a MOSFET may conduct in both directions between its drain and its source when it is on and may allow current to flow from its source to its drain (typically considered the reverse direction) when it is off. Diode 547 is installed in series with switch 546 to prevent the flow of current in the reverse direction (i.e. from source to drain) of switch 546.

[0105] Reference is made to FIG. 9, which illustrates the current flowing in switch 546. If switch 546 is closed at a time t₆, LC resonant tank 570 causes a substantially sinusoidal pulse 576 to be produced at the cathode of diode 548. Sinusoidal pulse 576 begins at about time t₆ and ends at about time t₇, and lasts for a time period t₈. The length of time period t₈ depends on the values of inductor 572 and capacitor 574. Time t₆ may be determined in known manner by calculation or experimentation. At time t₆ and again at time t₈, the current in switch 546 is zero, and accordingly, by closing switch 546 at time t₆ and then opening switch 546 at time t₇, the EMI emissions and switching losses of switch 546 may be substantially reduced.

[0106] The constant on-time of PDM control signal 542 is selected to exceed time period t₈, so that when switch 546 is opened at time t₇ (or later), switch 546 will essentially open a circuit which is carrying no current. During pulse 576, capacitor 574 will have become charged, and when switch 546 is opened, capacitor 574 will have no voltage across it but will have a charge on it. After switch 546 is opened, this charge is discharged into filter 550. The minimum off-time of PDM control signal 546 is selected to allow the charge on capacitor 574 to be essentially completely discharged. If the minimum off-time is too short, a charge will build up on capacitor 574 and the resonant operation of main power supply 518 will be degraded. The structure of PDM control circuit 516 is described below (FIG. 11).

[0107] Reference is again made to FIG. 8. As switch 546 opens and closes in response to PDM control signal 542, a series of pulses 576 are generated, forming a power signal V_(si-res). LC filter 550 smooths power signal V_(si-res) to produce power signal V_(s-res). The magnitude of power signal V_(s-res) during a particular time period will depend on the density of pulses 576 in power signal V_(si-res). Power signal V_(s-res) is diode-or'd with power signal V_(l) and V_(LV) by diodes 251 and 404 to form power signal V_(t), which is provided to power amplifier 204 at terminal 236.

[0108] Referring briefly to FIG. 2, PWM control signal 242 may have a duty cycle ranging from 0% to 100%. As a result, power signal V_(s) may range from 0 volts to the magnitude V_(max) of power source 212 (ignoring the negligible voltage drops across the components of main power supply 218).

[0109] Since PDM control signal 542 has a minimum off-time between each pulse 572 (FIG. 8), switch 546 cannot have a duty cycle of 100%. As a result, power signal V_(s-res) has a lower magnitude than the magnitude V_(max-res) of power supply 512 (which is analogous to power source 212). To allow power signal V_(s-res) to have the same magnitude as power signal V_(s) of the power amplifiers described above, the magnitude V_(max-res) of power supply 512 must be higher than the magnitude V_(max) of power source 212 (FIG. 2).

[0110] Since each pulse 576 produced by switch 546 will be identical, each pulse 576 will transfer a fixed amount of energy first into capacitor 574 and then into filter 550. The magnitude of power signal V_(s-res) will depend entirely of the density of pulses 576 (i.e. on the variable off-time between pulses). When a low power signal V_(s-res) is required, the density of pulses may be quite low and the frequency of the pulses may actually be in the audio band. Also, if the density of pulses is low, a large ripple may be seen in power signal V_(si-res), and if filter 550 has a desirable low time constant (which allows power signal V_(s-res) to more closely follow the output signal 232, as described above in relation to power signal V_(s) of power amplifier 400), power signal V_(s-res) may also have a corresponding large ripple.

[0111] In order to reduce the effects of this large ripple (which are described above in relation to power amplifier 400), low voltage power supply 402 is included in power amplifier 500, in the same manner as in power amplifier 400. The use of low voltage power rail 402 is not necessary, particularly where the magnitude of output signal 232 will be relatively high, and the disadvantages of high ripple (which are more disadvantageous at a low output level) will not be substantial. If low voltage power rail 402 is included in power amplifier 500, PDM control circuit 516 may be configured to disengage main power supply 518 when the magnitude of output signal 232 is expected to be less than threshold V_(d) (FIG. 7).

[0112] In this way, power signal V_(s) may be formed using PDM control circuit 542 and main power supply 518 while generating substantially less EMI than is generated by PWM control circuit 242 and main power supply 218.

[0113] Reference is next made to FIG. 10, which illustrates a seventh embodiment of a power amplifier 600 according to the present invention. Power amplifier 600 has an input terminal 202, an amplifier 204 and an output terminal 206 which are connected and operate in the same manner is in power amplifier 200. Power amplifier 600 has positive half circuit 608 and a negative half circuit 610, which is similar in structure and complementary in operation to positive half circuit 608. Positive half circuit 608 is similar to positive circuit 508 of power amplifier 500, with three exceptions. First, positive half circuit 608 does not include low voltage power supply 402, which was included in positive half circuit 508 and described above in relation to power amplifier 400. Second, positive half circuit 608 includes an overload detect block 362, the coupling and operation of which described above in relation to power amplifier 300. Third, positive half circuit 608 includes a rectifier 680, a summer 681, an amplifier 688, a summer 682 and a post regulator 684.

[0114] Rectifier 680 is coupled to offset block 211 to receive offset input signal 231 and to provide a rectified input signal 633, which corresponds to the power level required by amplifier 204 to produce output signal 232 in response to input signal 230. Rectified input signal 633 differs from rectified signal 240 in that no frequency compensation block (such as frequency compensation block 214) is used to produce rectified input signal 633. As a result, rectified input signal 633 does not have the phase lead effect of rectified signal 240 and the magnitude of rectified input signal 633 corresponds to the power required at a particular time, rather than a slightly later time.

[0115] Summer 681 is coupled to rectifier 680 to receive rectified input signal 633 and to overload detect block 362 (at a terminal 363) to receive overload detect signal 366. Summer 681 provides a regulation signal 690 equal to the difference between rectified input signal 633 and overload detect signal 366. Regulation signal 690 corresponds to rectified input signal 633 reduced by any excess power which could damage amplifier 204.

[0116] Amplifier 688 receives regulation signal 690 and provides an amplified regulation signal 692, which corresponds to regulation signal 690 but has a magnitude range which corresponds to the magnitude range of power signal V_(t).

[0117] Post regulator 684 may be a MOSFET type transistor, as shown in FIG. 9. The drain of post regulator 684 is coupled to the cathode of diode 251 to receive power signal V_(t). The source of post regulator 684 is coupled to positive power input terminal 236 to provide a regulated power signal V_(t-reg) to amplifier 204. The gate of post regulator 684 is coupled to summer 682. Summer 682 is coupled to amplifier 688 to receive amplified regulation signal 692 and to positive power input terminal 236 to receive power signal V_(t-reg). Summer 682 provides a regulator error signal 686 which is equal to the difference between amplified rectified input signal 683 and power signal V_(t-reg). Regulator error signal 686 reflects the difference between the power required by amplifier 204 to produce output signal 232 and the power actually being provided to amplifier 204 at positive power input terminal 236. If regulator error signal 686 is positive, then amplifier 204 is receiving more power than is required (or more power than is safe, based on overload detect signal 366). Post regulator 684 reduces power signal V_(t) received at the a drain of post regulator 684 and provides a smaller power signal V_(t-reg) to amplifier 204.

[0118] Normally, amplified regulation signal 692 will be configured to maintain post regulator 684 in a saturated condition so that power signal V_(t-reg) is essentially equal to power signal V_(t). The offset added by offset block 211 to input signal 230 to produce offset input signal 231 may be sufficient to produce this result. However, the amplification factor of amplifier 688 may also be used to ensure that post regulator 684 is normally saturated. However, when an overload condition is detected by overload detector 362, regulation signal 690 and amplified regulation signal 692 will fall quickly and post regulator 684 will act to limit power signal V_(t-reg). Post regulator 684 operates as a linear regulator, and preferably, it is highly responsive to changes in its gate voltage. Post regulator may be a power transistor, such as a power MOSFET, IGBT, BJT, or any other device which may be used a regulator.

[0119] In this way post regulator 684 is able to quickly reduce the power signal V_(t-reg) provided to amplifier 204 at positive power input terminal 236. Post regulator 684 is particularly useful when amplifier 204 is implemented as a power integrated circuit which might be damaged by an overload in microseconds.

[0120] As noted above, positive half circuit 608 does not include low voltage power supply 402. In addition to regulating the level of power supply V_(t-reg) when an overload occurs, post regulator 684 also smooths power signal V_(t) so that power signal V_(t-reg) has less ripple than power signal V_(t). As noted above in relation to power amplifier 400, one reason for using low voltage power supply 402 to eliminate the use of main power supply 218 (or resonant switching regulator 518) was to reduce eliminate the problem of a relatively large ripple on power signal V_(t) when power signal V_(t) had a relatively low magnitude. Since this ripple will be reduced by post regulator 684, the need for low voltage power supply 402 is reduced. If desired, low voltage power supply 402 may be incorporated into power amplifier 600 and a person skilled in the art will be capable of doing so.

[0121] Power amplifiers 200, 280, 300, 400, 500 and 600 utilize a feedback control system to ensure that power signal V_(t) (or V_(t-reg) in power amplifier 600) provides sufficient power to amplifier 204 to allow it to produce output signal 232 without excessive headroom. This feedback circuit will now be described, in the context of power amplifier 600, with reference to FIGS. 11 and 12.

[0122]FIG. 11 illustrates terminal 217 (FIG. 10), at which rectifier 215 produces rectified signal 240, summers 237 and 368, peak detector 220, differentiator 222, switch 224, part of PDM control circuit 516, a bias point network 702 and a rectified signal amplification network 704.

[0123] Bias point network 702 comprises resistor R₁, R₂, R₃ and a zener diode D₁. Resistor R₁ is coupled between ground and a terminal 706. Resistor R₂ is coupled between terminal 706 and a terminal 708. Resistor R₃ is coupled between terminal 708 and the most negative point of power amplifier 600 (FIG. 9). The most negative point of power amplifier 600 will, in general, be substantially more negative than the ground of power amplifier 600 and a person skilled in the art will be capable of selecting such a point. Zener diode D₁ is coupled between terminal 708 and ground. Resistors R₁ and R₂ form a voltage divider across zener diode D₁ so that terminal 706 will have a voltage less than ground (i.e. if zener diode D₁ has a zener voltage of 5 volts, R₁=10 kΩ and R₂=25 kΩ, then terminal 706 will have a voltage of −1.43 volts). The voltage at terminal 706 provides a bias voltage for transistors Q₂ and Q₃, which are described below. In general, the bias voltage will be set to be less than −0.7 volts.

[0124] Summers 237 and 368 are combined and implemented as a single summing network 708. Summing network 708 comprises transistors Q₁, Q₂, Q₃ and Q₄, resistors R₄, R₅, R₆ and R₇, a diode D₂ and a capacitor C₁. The emitter of pnp transistor Q₁ is coupled to terminal 236 through resistor R₄ to receive power signal V_(t-reg). Power signal V_(t-reg) is converted into a current signal i_(t) by resistor R₄. The base of transistor Q₁ is coupled to ground, and accordingly, a current signal i_(o) is provided at the collector of transistor Q₁. Since the base of transistor Q₁ is coupled to ground, current signal i_(o) will generally be equal current signal i_(t) (so long as power signal V_(t-reg) exceed ground by 0.7 volts). Current signal i_(o) represents the magnitude of output signal V_(t-reg) at any particular time.

[0125] The collector of transistor Q₁ is coupled to the collector of diode-connected transistor Q₃. The collector of transistor Q₃ is also coupled to terminal 363 through diode D₂ to receive overload detect signal 366, which is a current signal and is shown as current signal i_(d). The emitter of transistor Q₃ is coupled ground through a resistor R₆. Terminal 363 is coupled to ground through capacitor C₁ which operates to integrate and smooth overload detect signal 366. Diode D₂ ensures that no current flows into terminal 363 from summing network 708. Current signal i_(o) and overload detect signal i_(d) are summed at the collector of transistor Q₃ to form a current signal i_(sum), which flows from the collector to the emitter of transistor Q₃.

[0126] Transistor Q₄ is coupled to transistor Q₃ as a current mirror. The base of transistor Q₄ is coupled to the base and collector of transistor Q₃. The emitter of transistor Q₄ is coupled to ground through resistor R₇. A current equal to i_(sum), flows from the collector to the emitter of transistor Q₄. This current i_(sum) is drawn from a terminal 710.

[0127] The emitter of pnp transistor Q₂ is coupled to terminal 217 through resistor R₅ to receive rectified signal 240, which corresponds to the amount of power required by amplifier 204 to produce output signal 232 (FIG. 10). The base of transistor Q₂ is coupled to terminal 706 and the emitter of transistor Q₂ is coupled to terminal 710. Resistor R₅ converts rectified signal 240 into a current signal i_(r), which flows from the emitter to the collector of transistor Q₂, as long as rectified signal 240 exceeds the bias voltage at terminal 706 by 0.7 volts. Since rectified signal 240 is always above 0 volts, and since the bias voltage will generally be selected to less than −0.7 volts, current i_(r) will flow when ever rectified signal 240 is non-zero.

[0128] A current signal i_(e), flows out from terminal 710 to terminal 712. Current signal i_(e) will be equal to current i_(r) less current i_(sum). Current signal i_(e) is error signal 239 (FIGS. 1 and 9) and corresponds to the difference between the power that amplifier 204 requires and the power that it is presently receiving.

[0129] One part of PDM control signal 516 includes transistor Q₇. The base of transistor Q₇ is coupled to terminal 712. The collector of transistor Q₇ is coupled to a voltage source V_(cc) and the emitter of transistor Q₇ is coupled to terminal 243. Transistor Q₇ is coupled as an emitter-follower and simply buffers error signal 239 (or i_(e)) and provides control signal 244 (equal to error signal 239) at terminal 243.

[0130] Rectified signal amplification circuit 704 includes transistors Q₅ and Q₆ and resistors R₈ and R_(g). The emitter of pnp transistor Q₅ is coupled to terminal 217 through resistor R₈. The base of transistor Q₅ is coupled to terminal 706. The collector of transistor Q₅ is coupled to terminal 714. Resistor R₉ is coupled between terminal 714 and ground. Resistor R₉ is selected double the resistance of resistor R₈. The base of transistor Q₆ is coupled to terminal 714. The collector of transistor Q₆ is coupled to a voltage source V_(cc) and the emitter of transistor Q₆ is coupled to a terminal 716. Resistor R₈ and transistor Q₅ operate in the same manner as (and may actually be matched to) resistor R₅ and transistor Q₂ to provide a current signal i_(p) (which may be equal to current signal i_(r)) at the collector of transistor Q₅. Current signal i_(p) corresponds to the magnitude of rectified signal 240. Current signal i_(p) flows through resistor R₉ to ground, with the result that the voltage at terminal 714 is equal to 2 times the voltage of rectified signal 240 at any particular time. Transistor Q₇ is configured as an emitter-follower and simply buffers the voltage at terminal 714 to a voltage signal V_(rect) (equal to 2 times rectified signal 240) at terminal 716.

[0131] Peak detector 220 consists of a capacitor C₉ and a resistor R₁₀ coupled in series between terminal 716 and ground. Peak detector 220 operates in known manner to provide peak signal 252 at terminal 718.

[0132] Differentiator 222 consists of a capacitor C₃ and resistor R₁₁ and R₁₂. Capacitor C₃ and resistor R₁₁ are coupled in series between terminal 718 and ground. Resistor R₁₂ is coupled between the junction of capacitor C₃ and resistor R₁₁ and a terminal 720. Differentiator 222 operates in known manner to provide differential signal 254 at terminal 720.

[0133] Switch 224 comprises transistors Q₈ and Q₉ and resistors R₁₃, R₁₄ and R₁₅. The base-emitter junction of transistor Q₈ is coupled between terminal 720 and ground. The collector of transistor Q₈ is coupled to the base of transistor Q₉ through resistor R₁₄. Resistor R₁₄ and the base of transistor Q₉ are coupled to voltage source V_(cc) through resistor R₁₃. The emitter of pnp transistor Q₉ forms terminal b of switch 224 and is coupled to terminal 243 to receive control signal 244. The collector of transistor Q₉ forms terminal a of switch 224. Terminal a is coupled to the gate of linear regulator 226 (FIGS. 3 and 10). The collector of transistor Q₉ is coupled to ground through resistor R₁₅. The ground side connection of resistor R₁₅ forms terminal c of switch 224.

[0134] Switch 224 operates as follows. When differentiated signal 254 exceeds 0.7 volts, the base-emitter junction of transistor Q₈ is forward biased, and transistor Q₈ turns on. Resistor R₁₃ and R₁₄ form a voltage divider between voltage source V_(cc) and ground, thereby setting a base voltage for pnp transistor Q₉. When control signal 244 (at the collector of transistor Q₉) exceeds this base voltage, transistor Q₉ conducts and couples terminal 243 (and terminal b) to terminal a, thereby allowing control signal 244 to control linear regulator 226. If differentiated signal 254 does not exceeds 0.7 volts, then transistor Q₈ does not conduct and the base of transistor Q₉ is pulled up to V_(cc) through resistor R₁₃. Voltage source V_(cc) is selected so that control signal 244 will not exceed it and therefore, transistor Q₉ will not conduct. Terminal a will be pulled down to ground through resistor R₁₅, and will be effectively coupled to terminal c. Threshold 256 is considered to be exceeded if both (i) differentiated signal 254 exceeds 0.7 volts and (ii) control signal 244 exceeds the base voltage of Q₉ when condition (i) is met.

[0135] In this way, rectified signal 240, power signal V_(t-reg) and overload detect signal 366 are used to generate error signal 239 and control signal 244. Switch 244 operates in response to differential signal 254 to selectively couple control signal 244 to the gate of linear regulator 226 (FIGS. 3 and 10).

[0136]FIG. 12 illustrates another portion of PDM control circuit 516 which uses error signal 239 to generate PDM control signal 542. PDM control circuit 516 comprises AND gates G₁ and G₂, buffers B₁ and B₂, resistors R₁₆, R₁₇ and R₁₈, capacitors C₄, C₅ and C₆ and diodes D₃ and D₄.

[0137] One input terminal of AND gate G₁ is coupled to terminal 712 to receive error signal 239. The other input terminal of AND gate G₁ is coupled to node 722, to receive a delay signal 724, the operation of which is explained below. When both error signal 239 and delay signal 724 exceed the high input threshold 721 of AND gate G₁, AND gate G₁ provides a high signal to the clock inputs CLK of buffers B₁ and B₂.

[0138] Buffer B₁ has a DATA input which is coupled to voltage source V_(cc) (which is a high signal). When AND gate G₁ provides a high signal to the clock input CLK of buffer B₁, buffer B₁ provides a high signal at its Q output and at a node 728. At the same time, buffer B₁ will provide a low signal at its Q-not output. The Q output of buffer B₁ is coupled to ground through resistor R₁₆ and capacitor C₄. The junction of resistor R₁₆ and capacitor C₄ forms a terminal 726, which is coupled to a RESET terminal of buffer B₁. Diode D₃ is coupled in parallel with resistor R₁₆ between nodes 726 and 728. Resistor R₁₆ and capacitor C₄ act a delay circuit. Assuming that capacitor C₄ is initially discharged (i.e. the signal at terminal 726 is low), the high output signal at the Q output of buffer B₁ will charge capacitor C₄. The rate at which capacitor C₄ is charged will depend on the time constant of resistor R₁₆ and capacitor C₄. Eventually the voltage at terminal 726 (and at the RESET input of buffer B₁) will become high, and buffer B₁ will reset is Q output and terminal 728 to a low signal and its Q-not output to a high signal. Capacitor C₄ will be discharged through diode D₃ and return to its initial discharged condition. Referring also to FIG. 12, buffer B₁ thereby produces a pulse 732 which remains high for a time period t₉ at node 728.

[0139] In an identical fashion, buffer B₂ produces a pulse 734 which remains high for a time t₁₀ at terminal 730. Time t₁₀ of pulse 734 will depend on the time constant of resistor R₁₇ and capacitor C₅. Diode D₅ provides a discharge path for capacitor C₅ when the Q output of buffer B₂ is low. In PDM control circuit 516, the values of resistors R₁₆ and R₁₇ and capacitors C₄ and C₅ are selected so that the time constant of resistor R₁₇ and capacitor C₅ is shorter than the time constant of resistor R₁₆ and capacitor C₄. As a result, pulse 734 is shorter than pulse 732 (i.e. time t₁₀ is shorter than time t₉).

[0140] The Q-not output of buffer B₁ is coupled to ground through resistor R₁₈ and capacitor C₆. The junction of resistor R₁₈ and capacitor C₆ forms node 722, which is coupled to the second input of AND gate G₁. When the Q-not output of buffer B₁ becomes high (at the end of pulse 732), resistor R₁₈ and capacitor C₆ acts a delay circuit. After a time t₁₁, capacitor C₆ will be sufficiently charged so that node 722 is a high signal. After this time, AND gate will re-initiate the again provide a high signal to the clock inputs CLK of buffers B₁ and B₂ when error signal 239 is a high signal. This may occur immediately or may occur after some delay.

[0141] The inputs of AND gate G₂ are coupled to terminals 728 and 730 to receive pulses 732 and 734. The output of AND gate G₂ is coupled to terminal 241 to provide PDM control signal 542. As described earlier, PDM control signal 542 regulates the power signal V_(s-res) produced by main power supply 518. PDM control consists of a series of pulses which have a constant on-time (during which switch 546 (FIG. 10) is closed, a minimum off-time following each pulse following each pulse during which switch 546 must remain open and a variable off-time between pulses which must exceed the minimum off-time and during which switch 546 remains open.

[0142] Reference is made to FIG. 13. The constant on-time, minimum on-time and variable on-time of PDM control signal 542 are configured as follows. Prior to time t_(a), the outputs of both buffers B₁ and B₂ are low (i.e. PDM control signal 542 is low). At this time, node 722 will have a high signal level.

[0143] At time t₂, error signal 239 exceeds the high input threshold 721 of AND gate G₁, which will then initiate the generation of pulses 732 and 734. At the same time, node 722 will fall to a low value. AND gate G₂ will begin a high pulse 740 on PDM control signal 542, since both of its inputs (at terminals 728 and 730) are high. High pulse 740 will end after time t₁₀, when pulse 734 ends. Since the time t₁₀ of pulse 734 is defined only by the time constant of resistor R₁₇ and capacitor C₅, this time constant fixes the constant on-time of each high pulse of PDM control signal 542 to be equal to time t₁₀.

[0144] Pulse 732 will continue until time t₉ has elapsed. During this period, node 722 has a low signal level, and therefore, the output of AND gate G₁ is low. As a result a new high pulse cannot begin on PDM control signal 542 until after time t₉. This defines the minimum off-time of PDM control signal 542 to be equal to t₉-t₁₀.

[0145] When pulse 732 ends after time t₉, the signal level of node 722 will rise and eventually will become a high signal again. After this time, a new high pulse will be initiated on PDM control signal 542 whenever error signal 239 exceeds the high input threshold 721 of AND gate G₁. This may occur immediately, as shown at 742 or after a delay, as shown at 744.

[0146] The circuits of FIG. 11 and 12 provide PDM control signal 542 and control signal 244, which control linear regulator 226 and main power supply 518 of power amplifier 600 (FIG. 9). The circuit of FIG. 10 may be modified to provide an error signal suitable for use in power amplifier 500 by removing the coupling between the collector of transistor Q₃ and terminal 363 (since power amplifier 500 does not include an overload detection block 362). The circuits of FIGS. 11 and 12 may be modified to provide a PWM control signal 242 to control the main power supply 218 of power amplifiers 200, 280, 300 or 400. A person skilled in the art will capable of making these amendments.

[0147] Reference is next made to FIG. 14, which illustrates an eighth embodiment of a power amplifier 700 according to the present invention. Power amplifier 700 is a bridge amplifier which can amplify two input signals 730 a and 730 b simultaneously. Like power amplifier 300 (FIG. 4), components of power amplifier 700 which relate only to input signal 730 a are identified by a reference numeral containing the letter “a” and components of power amplifier 700 which relate only to input signal 730 b are identified by a reference numeral containing the letter “b”. Power amplifier 700 has two input terminals 702 a and 702 b, a power supply circuit 708 and two bridge amplifiers 709 a and 709 b.

[0148] Bridge amplifier 709 a includes two amplifiers 704 a+ and 704 a− and an inverter 705 a. The input terminal of amplifier 704 a+ is coupled to input terminal 702 a to receive an input signal 730. The input terminal of amplifier 704 a− is coupled to input terminal 702 a through inverter 705 a. Amplifier 704 a+ amplifies positive half waves of input signal 730 a. Amplifier 704 a− amplifies negative half waves of input signal 730 a. Together, amplifier 704 a+ and 704 a− provide an output signal 732 a corresponding to input signal 730 a. Load 734 a is coupled between the outputs of amplifiers 704 a+ and 704 a− to receive output signal 732, which is formed between the output terminals of amplifiers 704 a+ and 704 a−.

[0149] Similarly, bridge amplifier 709 b includes two amplifiers 704 b+ and 704 b− and an inverter 705 b, which are coupled to input terminal 702 b to receive an input signal 730 b. Amplifiers 704 b+ and 704 b− cooperate to produce an output signal 732 b between their output terminals, to which load 734 b is coupled.

[0150] The detailed structure and operation of bridge amplifiers 709 a and 709 b are set out in U.S. Pat. No. 5,075,634, entitled COMPOSITE BRIDGE AMPLIFIER, which is incorporated herein by this reference.

[0151] Power supply circuit 708 is similar in structure and operation to the positive half circuits 208, 308, 408, 508 and 608. The primary difference between power supply circuit 708 and these positive half circuits is that power supply circuit 708 provides power to bridge amplifiers 709 a and 709 b during positive and negative half waves of input signals 730 a and 730 b. Accordingly there is no “negative half circuit” in power amplifier 700.

[0152] Power supply circuit 708 has two input signal compensation blocks 705 a and 705 b, which respectively comprise offset blocks 311 a and 311 b, frequency compensation blocks 314 a and 315 b and rectifiers 715 a and 715 b. In order to make power supply circuit 708 operative during positive and negative half waves of input signals 730 a and 730 b, rectifiers 315 a and 315 b of power amplifier 300 have been replaced with rectifiers 715 a and 715 b, which are full wave rectifiers and produce full wave rectified signals 740 a and 740 b which are diode-or'd by diodes 360 a and 360 b to produce a full wave rectified signal 740 at terminal 341.

[0153] Power supply circuit 708 also incorporates a number of other features of the power amplifiers described above: low voltage power supply 402, explained above in relation to power amplifier 400; overload detect blocks 362 a and 362 b, which respectively protect bridge amplifiers 709 a and 709 b and which were described above in relation to power amplifier 300; PDM control circuit 516 and main power supply 518, described above in relation to power amplifier 500; and post regulator 684, which was described above in relation to power amplifier 600.

[0154] Except as described above, power supply circuit 708 operates in a matter analogous to the positive half circuits described earlier and provides a power signal V_(t-reg) at terminal 736 from which bridge amplifiers 709 a and 709 b receive power. Power signal V_(t-reg) is sufficient to power both bridge amplifiers 709 a and 709 b.

[0155] Power amplifier 700 provides the advantages of reduced headroom and reduced EMI emissions which result from the use of resonant switching regulator 516 and low voltage power supply 402. Furthermore, power amplifier 700 does not require two half circuits to provide power to its amplifiers 204 a +, 204 a+, 204 b+ and 204 b−. Power amplifier 700 provides the advantage of a predictive control system for main power supply 518 and transient power supply 223 described above in relation to power amplifier 200.

[0156] Although power amplifiers 200, 280, 300, 400, 500, 600 and 700 may have reduced EMI emissions in comparison to prior art designs, a person skilled in the art will recognize that it is impossible to entirely eliminate EMI generation within such a device. Any EMI which is created within the power amplifier, and particularly within the positive and negative half circuits (or, in the case of bridge amplifier implementation, the power supply circuit) of the power amplifier may be coupled to the input terminal 202 of the amplifier. In addition, this EMI may be coupled on input signal 230 when it is received by power amplifier 200. Accordingly, it is desirable to (i) reduce the coupling of any EMI generated within the power amplifier to the input terminal and (ii) to reduce the coupling of any such EMI onto input signal 230 within the power amplifier.

[0157] Reference is next made to FIG. 15, which illustrates an EMI isolation circuit 800 which may be used to decouple EMI generated within the positive and negative half circuits of power amplifier 200 from the input signal 230. EMI isolation circuit 800 is explained here in the context of power amplifier 200, EMI isolation circuit 800 may also be used with power amplifiers 280, 300, 400, 500, 600 and 700.

[0158] EMI isolation circuit 800 consists of a non-inverting amplification amplifier 804 and an inverting reduction amplifier 806 coupled in series between an input terminal 802 and terminal 202. (In other embodiments of EMI isolation circuit, amplifiers 804 and 806 may be non-inverting or inverting, depending on their desired operation.) Terminal 202 is the same terminal as input terminal 202 of the power amplifier discussed above, however, input signal 230 is now received at an input terminal 802, which consists of a signal input terminal 802 a and a ground input terminal 802 b, and is coupled to terminal 202 through EMI isolation circuit 800. Amplifier 804 is coupled to input terminal 802 a to receive input signal 230.

[0159] Positive half circuit 208, negative half circuit 210 and EMI isolation circuit 800 will typically be encased in an EMI shield 810. EMI shield 810 may be the chassis of an enclosure in which power amplifier 200 is installed. Typically, EMI shield 810 will have a ground level which may be referred to as a chassis ground 812. Positive half circuit 208 and negative half circuit 210 will have a separate power amplifier ground 814, which is floating in relation to chassis ground 812. Ground input terminal 802 b and load 234 are coupled to chassis ground 812.

[0160] Reference numeral 816 identifies a coupling network between chassis ground 812 and power amplifier ground 814. Coupling network 816 actually comprises all of the paths between the two separate grounds.

[0161] Amplifier circuit 804 comprises an op-amp 820, resistors 822, 824 and 826 and a capacitor 828. The negative input terminal of op-amp 820 is coupled to chassis ground 812 through resistor 822 and to the output of op-amp 820 through resistor 824. The positive input terminal of op-amp 820 is coupled to signal input terminal 802 a through resistor 826. Signal input terminal 802 a is coupled to chassis ground 812 through capacitor 828. Resistors 822 and 826 are selected to have a large and equal resistance. Resistor 824 is selected to have a resistance much larger than that of resistor 822, thereby forming amplifier 804 into a non-inverting amplifying amplifier. Typically, resistors 822 and 826 may have a resistance of 100 kΩ and resistor 824 may have a resistance of 1 MΩ. With these values, amplifier 804 will operate as a multiply-by-10 amplifier. Preferably, amplifier 804 is configured to have an amplification of 2 to 20 times, and more preferably it will have a gain of 3 to 15 times.

[0162] Amplifier circuit 806 comprises an op-amp 830 and resistors 832, 834 and 836. The negative input of op-amp 830 is coupled to the output of op-amp 820 through resistor 832 and to the output of op-amp 830 through resistor 834. The positive input terminal of op-amp 830 is coupled directly to power amplifier ground 814 and to chassis ground through resistor 836. Resistor 836 forms part of coupling network 816. Typically, resistor 836 will have a small resistance such as 1 kΩ. Resistor 832 is selected to have a much larger resistance than resistor 834, forming amplifier circuit 806 into an inverting reducting amplifier. Typically, resistor 832 may have a resistance of 10 kΩ and resistor 834 may have a resistance of 1 kΩ. With these values, amplification circuit 806 will act as a divide-by-10 reducing amplifier.

[0163] EMI isolation circuit 800 reduces the coupling of EMI generated within positive half circuit 208 and negative half circuit 210 onto input signal 230 within power amplifier 230 as follows. Within the context of EMI isolation circuit 800, any such EMI may be seen as an EMI signal 835 across resistor 836, which is the only coupling between the floating power amplifier ground 814 and the chassis ground 812. Input signal 230 is received across terminal 802 a and 802 b. Input signal 230 is amplified by amplifier 804, which provides an amplified input signal 842 across nodes 840 a and 840 b corresponding to input signal 230. Input signal 230 combined with EMI signal 835 form an EMI contaminated input signal 846 across terminal 844 a and 844 b. EMI contaminated input signal 846 is reduced by amplifier 806, providing an EMI-decoupled input 830 at terminal 202. This EMI-decoupled input signal 830 is then amplified by power amplifier 200, as described above to produce output signal 232.

[0164] EMI-decoupled input signal 830 will correspond substantially to input signal 230 with a relatively small degree of contamination from EMI signal 836. This effect may be seen through the following example. If input signal 230 has a magnitude of 3 volts, EMI signal has a magnitude of −1, amplifier 804 has an amplification of 10 and amplifier 806 has an amplification of 0.1, then amplified input signal 842 will have a magnitude of 30, EMI contaminated input signal 846 will have a magnitude of 29 and EMI-decoupled input signal 830 will have a magnitude 2.9. By amplifying input signal 230 by a selected factor before it is contaminated by EMI signal 835 and then reducing EMI contaminated input signal 846 by the same factor, the effect of EMI signal 835 on input signal 230 is reduced by the selected factor, and consequently, the effect of EMI signal 835 on the operation of power amplifier 200 is reduced. It is not necessary that the amplification factor of amplifier 806 be the reciprocal of the amplification factor of amplifier 806. The amplification factors of amplifiers 804 and 806 may be varied to provide a desired degree of reduction of EMI signal 835 and an appropriate input signal for power amplifier 200.

[0165] EMI isolation circuit 800 also operates to decouple EMI signal 835 from input terminal 802, and thereby reduces the effect of EMI signal 835 on output signal 232. An EMI signal 850 corresponding to EMI signal 835 will be injected into terminals 849 a and 849 b at the positive and negative inputs of op-amp 820. Resistors 822 and 826 have a high resistance, as described above, and provide a high impedance to EMI signal 850, thereby substantially decoupling EMI image signal 850 from input terminal 802. Amplifier 804, which receives input signal 230 from input terminal 802 thus receives an input signal which is not substantially contaminated by EMI.

[0166] By using EMI isolation circuit 800 as an input circuit coupled to the input terminal 202 of a power amplifier according to the present invention, EMI generated within the power amplifier may be substantially decoupled both from the input terminal 202 and the input signal 230.

[0167] Reference is next made to FIG. 16, which is a block diagram of a ninth power amplifier 900 according to the present invention. Power amplifier 900 has an input terminal 102, an amplifier 104, and output terminal 106, a positive half circuit 908 and a negative half circuit 910, which has the same structure and a complementary operation to that of positive half circuit 908. In power amplifier 900, input compensation block 105, transient detect block 119, control circuit 116, main power supply 118 and amplifier 104 operate as in power amplifier 100. The transient operation of power amplifier 900 differs from that of power amplifier 100. Transient power supply 123 has been replaced by first and second transient control circuits 923 a and 923 b, which operate in conjunction with summer 937 and OR gate 902 to modify the power signal V_(s) produced by main power supply 118.

[0168] Transient control circuit 923 a has two digital output signals: a first transient control signal 904 and a second transient control signal 908. In this exemplary embodiment of power amplifier 900, transient control signal 904 is normally low and transient control signal 908 is normally high when power amplifier 900 is in its normal operation.

[0169] When transient detect block 119 detects a change in input signal 130 that requires power amplifier 900 to enter transient operation, transient control circuit 923 a generates a high pulse in transient control signal 904, which is OR'd with control signal 942 to produce a main power supply control signal 906. Main power supply control signal 906 is high when either control signal 142 or first transient control signal 904 is high. Control signal 942 is a PWM signal similar to control signal 242 of power amplifier 200, and can have a duty cycle between 0 and 100%, depending on the value of compensated input signal 140. Main power supply 118 is responsive to main power supply control signal 906 and produces power signal V_(s) with a magnitude corresponding to main power supply control signal 906. When transient control signal 904 is low, main power supply control signal 906 is identical to control signal 142, and main power supply 118 provides power signal V_(s) as in power amplifier 100. When transient control signal 904 is high, main power supply control signal 906 will be high (i.e. it will have a duty cycle of 100%), and main power supply 118 will provide power signal V_(s) at its highest voltage.

[0170] When transient detect block 119 detects a change in input signal 130 that requires power amplifier 900 to enter transient operation, transient control circuit 923 a also generates a low pulse on transient control signal 908. Transient control circuit 923 b receives the low pulse and increases the magnitude of error signal 939 in response to it. Control circuit 116 receives the increased error signal and increases the duty cycle of control signal 142 in response. After a selected time, transient control circuit 923 a ends the high pulse on transient control signal 904 and main power supply 118 becomes responsive to control signal 142. Main power supply 118 will provide power signal V_(s) with a voltage level corresponding to the increased duty cycle of control signal 142.

[0171] Transient control circuit 923 b includes a fast attack block 910 and a slow release block 912. Fast attack block 910 operates to quickly increase the magnitude of error signal 939 in response a low pulse on transient control signal 908. Slow release block 912 operates to slowly reduce the increase in error signal 939, until, after a selected time, transient control circuit 923 b has no effect on error signal 939. Power amplifier 900 then returns to normal operation.

[0172] Power amplifier 900 does not have a transient power supply (such as transient power supply 123 or 223). Instead, first and second transient control circuits 923 a and 923 b modify the power signal V_(s) provided by main power supply by increasing the duty cycle of main power supply control signal 906 when power amplifier 900 is in its transient operation.

[0173] Reference is next made to FIG. 17, which illustrates a power amplifier 1000. Power amplifier 1000 is a more detailed embodiment of a power amplifier based on the general structure of power amplifier 900. Power amplifier 1000 has an input terminal 202, an amplifier 204, an output terminal 206, a positive half circuit 1008 and a negative half circuit 1010. Negative half circuit 1010 has the same structure and a complementary operation to that of positive half circuit 1008. Positive half circuit 1008 includes an input signal compensation block 205, a transient detect block 219, a control circuit 216 and a main power supply 218, which are coupled and operate in the same manner as the corresponding components in power supply 200 (FIG. 2). Positive half circuit also includes first and second transient control circuits 1023 a and 1023 b and an OR gate 1002, which correspond to first and second transient control circuits 923 a and 923 b and OR gate 902 of power amplifier 900 (FIG. 16).

[0174] First transient control circuit 1023 includes two one-shot circuits 1030 and 1032, both of which are responsive to transient signal 257. One-shot circuit 1030 provides transient control signal 1004, which corresponds to transient control signal 904 of power amplifier 900 (FIG. 16). When power amplifier 1000 in its normal operation, transient control signal 1004 is low. One-shot circuit 1032 provides transient control signal 1008, which corresponds to transient control signal 908 of power amplifier 900 (FIG. 16). When power amplifier 1000 is in its normal operation, transient control signal 1008 is high. First transient control signal 1004 is OR'd with control signal 142 to generate main power supply control signal 1006, which controls switch 246 of main power supply 218.

[0175] When transient signal 257 becomes high, power amplifier 1000 enters its transient operation. In response to transient signal 257 becoming high, one-shot circuit 1030 provides a high pulse 1034 on transient control signal 904, causing main power supply control signal 1006 to become high. Switch 246 of main power supply 218 remains closed while transient control signal 904 is high, and main power supply 218 provides power signal V_(s) at its maximum voltage.

[0176] Transient control circuit 1023 b comprises diodes 1038, 1040, capacitors 1042, 1044 and 1046, resistors 1048 and 1050 and an amplifier 1052. The cathode of diode 1038 is coupled to the output of one-shot circuit 1032. The anode of diode 1038 defines a node 1054 and is coupled to ground through capacitor 1042. Node 1054 is coupled to the cathode of diode 1040 through parallel connected resistor 1048 and capacitor 1044. The anode of diode 1040 is coupled to a node 1056, which is also coupled to the output of amplifier 235 and to summer 237. Amplifier 1052 is coupled between the output of summer 237 and a node 1058, where it provides an amplified error signal 1039 which corresponds to error signal 239. Resistor 1050 and capacitor 1046 are coupled in parallel between nodes 1056 and 1058. Control circuit 216 is coupled to terminal 1058 and is responsive to amplified control signal 1039 to produce control signal 242.

[0177] The operation of power amplifier 1000 will now be explained with reference to FIG. 18. Prior to time t₁₂, power amplifier 1000 is in its normal operation. Capacitor 1042 is charged so that node 1054 has a voltage V₁₀₅₄ equal to or higher than the voltage or error signal V₁₀₅₆ at node 1056. Transient control signal 1004 is low and transient control signal 1008 is high. The normal level of transient control signal 1008 is selected to be approximately equal to or slightly higher than the maximum voltage V₁₀₅₆ (i.e. the maximum voltage across capacitor 1042 when it is fully charged). Control circuit 216 generates control signal 242 in response to amplified error signal 1039 at terminal 1058. Main power supply control signal 1006 is identical to control signal 242.

[0178] Diode 1040 is optional and may be provided to prevent current from flowing from node 1054 to node 1056 when capacitor 1042 is charged. Amplifier 1000 (like amplifiers 100, 200, 280, 300, 400, 500, 600, 700 and 900) uses a closed loop feedback through amplifier 235, summer 237 and control circuit 216 to reduce error signal 239 by keeping voltage V₁₀₅₆ approximately equal to the voltage V₁₀₆₀ of 1060 (to the extent possible, given the limits of main power supply 218). Amplifier 1052, resistor 1050 and 1046 provide an additional feedback loop, to enhance the control of error signal 239 and to keep voltages V₁₀₅₆ and V₁₀₆₀ approximately equal. Amplifier 1052, resistor 1050 and 1046 may be optionally provided in any of amplifiers described above. In normal operation, capacitor 1044 will be discharged.

[0179] At time t₁₂, transient detect block 219 detects a large transient in the level of input signal 230 and sets transient signal 257 high. In response, one-shot circuit 1030 sets transient control signal 1004 high for a selected time period t₁₃ and one-shot circuit 1032 sets transient control signal 1008 low for a selected time period t₁₄. Capacitor 1042 begins to discharge through diode 1038 into one-shot circuit 1032, causing voltage V₁₀₅₄ to fall. The low voltage of transient control signal 1008 is selected to sufficiently discharge capacitor 1042 that the one-shot begins to draw current from node 1056. For example, the low voltage of transient control signal 1008 may be selected to be ground, so that capacitor 1042 is fully discharged (assuming that the length of the low pulse is sufficiently long).

[0180] As voltage V₁₀₅₄ falls, the feedback structure of power amplifier 1000 attempts to keep voltage V₁₀₅₆ approximately equal to voltage V₁₀₆₀. The feedback operation and amplifier 1052 force the magnitude of error signal 239, and the amplified error signal 1039 to rise. Resistors 1048 and 1050 act as a voltage divider. Since voltage V₁₀₅₆ is controlled by the feedback structure, and since voltage V₁₀₅₄ is falling as capacitor 1042 discharges, the voltage divider helps control the voltage to which amplified error signal 1039 rises. The rate at which error signal 239 and amplified error signal 1039 rise, during an initial period t₁₅, depends on the relative resistances of resistors 1048 and 1050 and on the rate at which voltage V₁₀₅₄ falls, which in turn depends on the rate at which capacitor 1042 is discharged by one-shot circuit 1032. As amplified error signal 1039 rises, control circuit 216 increases the duty cycle of control signal 242.

[0181] As capacitor 1042 is discharged, one-shot circuit 1032 begins to draw current from node 1056, and from node 1058. Capacitors 1044 and 1046 begin to charge, allowing error signal 239 and amplified error signal 1039 to initially fall and then vary depending on the value of V_(targ), which was described above, during time period t₁₆.

[0182] During time period t₁₃, main power supply control signal 1006 is high, causing main power supply to increase the magnitude of power supply V_(s) through filter 250. During this time, the power signal V_(t), which is identical to V_(s), supplied to amplifier 204 increases, taking into account the relatively slow time constant of filter 250. Since switch 246 is held closed during this period, power signal V_(t) will rise relatively rapidly and amplifier 204 will receive sufficient power to produce output signal 232 in response to input signal 230.

[0183] When time period t₁₃ ends, transient control signal 1004 becomes low and main power supply control signal 1006 again follows control signal 242. Time period t₁₃ is selected to be sufficiently long that (i) the magnitude of amplified error signal 1039 has risen, as described above in relation to time period t₁₅ and the early part of time period t₁₆ and (2) that the duty cycle of control signal 242 has increased sufficiently that main power supply 218 will provide sufficient power to amplifier 204 in response to control signal 242. Time period t₁₃ is preferably not substantially longer than is required for these conditions to be met, since any excess power delivered to amplifier 204 will be dissipated by amplifier 204.

[0184] After time period t₁₄, transient control signal 1008 becomes high again. Capacitor 1054 begins to charge until voltage V₁₀₅₄ is equal to or high than voltage V₁₀₅₆ (which will be a constantly varying voltage, depending on the magnitude of power signal V_(t). If voltage V₁₀₅₆ subsequently exceeds V₁₀₅₄, capacitor 1042 will again charge until voltage V₁₀₅₄ is equal to or higher than voltage V₁₀₅₆.

[0185] During time t₁₆, amplified error signal 1039 remains amplified by the feedback operation described above and by the voltage divider formed by resistors 1048 and 1050. At time t₁₇, power amplifier 1000 returns to normal operation and amplified error signal 1039 returns gradually to its normal value, which will depend on the difference between rectified signal 240 and compensated power signal V_(tr).

[0186] The length of time period t₁₄ is selected to be long enough to allow filter 250 to charge sufficiently that power signal V_(s) is larger than V_(req), and more preferably has reached V_(targ). This will depend on the time constant of filter 250 in any particular embodiment of a power amplifier according to the present invention.

[0187] Power amplifier 1000 responds to transient signal 257 by increasing the duty cycle of main power supply control signal 1039, rather than by engaging a separate transient power supply, such as transient power supply 223 of power amplifier 200 (FIG. 2). Since power amplifier 1000 has only one power supply (main power supply 218), there is no need for diode 251 (FIG. 2).

[0188] The structure of power amplifiers 900 and 1000 may be combined with the overload detect blocks 362 of power amplifier 300 (FIG. 4) and may be used to power two or more channels, as described in relation to power amplifier 300. Power amplifiers 900 and 1000 may also be provided with low voltage power supplies 402 (and diodes 404) as described in relation to power amplifier 400 (FIG. 6). Power amplifiers 900 and 1000 may also be provided with a main power supply incorporating a resonant switching regulator, such as main power supply 518 described in relation to power amplifier 500 (FIG. 8), in place of main power supply 218. Power amplifier 900 and 1000 may also be provided with a post regulator 684 and the associated circuitry (rectifier 680, summer 681, amplifier 688 and summer 682), as described in relation to power amplifier 600 (FIG. 10). Power amplifier 900 and 1000 may also be implemented as a bridge amplifier by modifying rectifier 215 to be a full wave rectifier as described in relation to power amplifier 700 (FIG. 14). The EMI isolation circuit 800 of FIG. 15 may be used to decouple EMI from the input terminal 202.

[0189] A number of preferred embodiments of the present invention have been described. Many variations may be made on these embodiments, and in particular, features of some embodiments may be incorporated into other embodiments to provide an embodiment suited for a particular application. Such variations and other will be obvious to persons skilled in the art and will fall within the scope of the present invention, which is limited only by the appended claims. 

We claim:
 1. A power amplifier for receiving a first input signal at a first input terminal and for producing a first output signal at a first output terminal, said first output signal corresponding to said first input signal, a first signal amplifier being coupled to the input terminal to receive the input signal and coupled to the output terminal to provide the output signal, the first signal amplifier having a first power terminal for receiving a total power signal and said power amplifier having a first power supply circuit comprising: (a) a first input signal compensation block coupled to the first input terminal to receive the first input signal and to provide a compensated input signal corresponding to the first input signal, wherein the compensated input signal defines a target power level; (b) a power signal compensation block for receiving the total power signal and for providing a compensated power signal corresponding to the total power signal; (c) a summer coupled to the input signal compensation block and to the power signal compensation block for providing an error signal corresponding to a difference between the target power level and the power level of the total power signal; (d) a control circuit coupled to the summer for receiving the error signal and for providing a first control signal and a second control signal in response to the error signal, wherein the first control signal corresponds to a target main power signal level and the second control signal corresponds to a target transient power signal level; (e) a transient detect block coupled to the input signal compensation block for providing a transient signal to identify a transient condition when the rate of change in the slew rate of the compensated input signal exceeds a selected transient threshold; (f) a main power supply for providing a main power signal at the first power terminal in response to the first control signal; and (g) a selectively engageable transient power supply for providing a transient power signal at the first power terminal in response to the second control signal and the transient signal, wherein the transient power supply is engaged when the transient signal indicates that a transient condition exists; wherein the control circuit provides the first and second control signals such that the target main power signal level is equal to or higher than the target transient power signal level and wherein the magnitude of the total power signal is generally equal to the higher of the magnitude of the main power signal or the magnitude of the transient power signal.
 2. The power amplifier of claim 1 wherein the first input compensation circuit includes: (i) an offset block for adding an offset to said input signal to provide an offset input signal; (ii) a frequency compensation block for receiving the offset input signal and for providing a corresponding frequency compensated circuit having its voltage components phase advanced with respect to its current component; and (iii) a first rectifier for rectifying the frequency compensated circuit to provide the compensated input signal.
 3. The power amplifier of claim 2 wherein the frequency compensation block is configured to amplify the amplitude of selected the frequency compensated signal at selected frequency components, wherein said selected frequency components exceed a selected frequency compensation threshold.
 4. The power amplifier of claim 3 wherein the amplitude of the selected frequency components is progressively amplified to a greater extent.
 5. The power amplifier of claim 3 wherein the amplitude of the selected frequency components is equally amplified.
 6. The power amplifier of claim 2 wherein the first rectifier is a half wave rectifier.
 7. The power amplifier of claim 2 wherein the input offset block is configured to add a smaller offset to the first input signal if the highest frequency component of the input signal is less than a selected offset frequency threshold and to add a larger offset to the first input signal otherwise.
 8. The power amplifier of claim 1 wherein the first input compensation block provides the compensated input signal corresponding to a target power level that exceeds the sum of the power required by the first amplifier to generate a first output signal corresponding to the first input signal and at least half of a ripple in the main power signal.
 9. The power amplifier of claim 1 wherein the first control circuit is a PWM signal having a fixed switching frequency.
 10. The power amplifier of claim 9 wherein the main power supply is a switching regulator including: (i) a main power source; (ii) a switch coupled to the power source and responsive to the first control signal to provide an unfiltered main power signal; and (ii) an integrating filter coupled to switch to provide the main power signal corresponding to the unfiltered power signal.
 11. The power amplifier of claim 10 wherein the switching frequency is selected to limit the EMI emitted by the main power supply to a selected maximum EMI limit.
 12. The power supply of claim 10 wherein the transient detect block includes: (i) a peak detector for providing a peak signal corresponding to the peak envelope of the compensated input signal; (ii) a differentiator coupled to the peak detector for providing a differentiated signal corresponding to the rate of change of the compensated input signal; and (iii) a comparator for comparing the differentiated signal with the transient threshold to provide the transient signal; and wherein the transient power supply includes: (iv) a transient power source; (v) a transient power regulator coupled to the control circuit for receiving the second control signal; and (iv) a transient supply switch for engaging the transient power regulator in response to the transient signal.
 13. The power amplifier of claim 12 wherein the main power source and transient power source are the same.
 14. The power amplifier of claim 12 wherein the transient power regulator is a linear regulator.
 15. The power amplifier of claim 14 wherein the transient power regulator includes a MOSFET.
 16. The power amplifier of claim 11 wherein a time constant of the integrating filter is selected to effectively smooth the main power signal compared to the unfiltered power signal.
 17. The power amplifier of claim 12 wherein a discharge rate of the peak detector is selected to correspond to a slew rate of the main power supply.
 18. The power amplifier of claim 1 further including: (i) an overload detect block coupled to the first signal amplifier to provide an overload signal corresponding to one or more overload conditions within the first signal amplifier; and (ii) means for combining the overload signal with the compensated input signal to provide an adjusted compensated input signal; wherein the error signal corresponds to a difference between the adjusted compensated input signal and the compensated power signal.
 19. The power amplifier of claim 18 wherein the means for combining is the summer.
 20. The power amplifier of claim 18 wherein the means for combining is a second summer.
 21. The power amplifier of claim 1 wherein a second signal amplifier is coupled to the first power terminal and a second input signal is received at a second input terminal and wherein the first input signal compensation block provides a first compensated input signal and further including: (i) a second input signal compensation for providing a second compensated input signal; and (ii) a combining circuit for combining the first and second compensated input signals to provide the compensated input signal having a magnitude corresponding to the higher magnitude of the first and second compensated input signals.
 22. The power amplifier of claim 21 wherein the combining circuit is a pair of diodes for diode OR'ing the first and second compensated input signals.
 23. The power amplifier of claim 2 wherein the first control signal is a pulse density modulated control signal and wherein the main power supply is a resonant switching power regulator.
 24. The power amplifier of claim 23 wherein the main power supply is a zero-current switching regulator and includes a LC resonant tank.
 25. The power amplifier of claim 1 further including a low voltage power supply coupled for providing a fixed low voltage power signal to the first power terminal, wherein the total power signal is generally equal to the higher of the magnitude of the main power signal, the magnitude of the transient power signal or the magnitude of the low voltage power signal.
 26. The power amplifier of claim 25 wherein the control circuit is configured to set the main power signal to zero when the target power level is less than the magnitude of the low voltage power signal.
 27. The power amplifier of claim 23 further including a post regulation circuit having: (i) an overload detect block coupled to the first signal amplifier to provide an overload signal corresponding to one or more overload conditions within the first signal amplifier; (ii) a second rectifier coupled to the offset block for receiving the offset input signal and providing a rectified input signal; (iii) a third summer for subtracting the overload signal from the rectified input signal to provide a regulation signal; (iv) a regulation amplifier coupled to the third summer for providing a amplified regulation signal corresponding to the regulation signal and having a magnitude range corresponding to the magnitude range of the total power signal; (v) a post regulator having a control terminal, and coupled between the main and transient power supplies and the first power terminal; and (vi) a regulation feedback circuit coupled between the first power terminal and the control terminal of the post regulator and including a fourth summer for providing a regulator error signal corresponding to the difference between the total power signal and the amplified regulation signal; wherein the post regulator regulates the total power signal in response to the regulator error signal when an overload condition occurs.
 28. The power amplifier of any one of claims 2-5 or 7-27 wherein the first amplifier is a bridge amplifier and wherein the first rectifier is a full wave rectifier.
 29. The power amplifier of any of claims 1-27 wherein the first signal amplifier has a second power terminal and further including a second power supply circuit having the same structure as said first power supply circuit, wherein said first power supply circuit supplies power to said first signal amplifier at said first power terminal during positive half wave of said output signal and said second power supply circuit provides power to said first signal amplifier at said second power terminal during negative half waves of said output signal.
 30. The power amplifier of claim 1 further including an EMI isolation circuit coupled between said first input terminal and an internal input terminal for providing a first EMI-decoupled signal corresponding to said first input signal at said internal input terminal, and wherein said first input signal compensation block and said first signal amplifier are coupled to said internal input node, wherein the EMI isolation circuit has: (i) a first isolation amplifier having a first amplification factor coupled to said first input terminal through a first impedance for receiving said input signal and for providing an amplified input signal; and (ii) a second isolation amplifier having a second amplification factor coupled to said first amplifier for receiving said amplified input signal and to said internal input terminal for providing said first EMI-decoupled signal; wherein the amplification factor of the first isolation amplifier is greater than 1 and the amplification factor of said second isolation amplifier is less than
 1. 31. The power amplifier of claim 30 wherein said first impedance is greater than 50 kΩ.
 32. The power amplifier of claim 30 wherein said first impedance is equal to or greater than 100 kΩ.
 33. The power amplifier of claim 30 wherein the EMI isolation circuit further has an EMI shield for encompassing the first and second isolation amplifiers and the first power supply circuit.
 34. A power amplifier for receiving a first input signal at a first input terminal and for producing a first output signal at a first output terminal, said first output signal corresponding to said first input signal, a first signal amplifier being coupled to the input terminal to receive the input signal and coupled to the output terminal to provide the output signal, the first signal amplifier having a first power terminal for receiving a total power signal and said power amplifier having a first power supply circuit comprising: (a) a first input signal compensation block coupled to the first input terminal to receive the first input signal and to provide a compensated input signal corresponding to the first input signal, wherein the compensated input signal defines a target power level; (b) a power signal compensation block for receiving the total power signal and for providing a compensated power signal corresponding to the total power signal; (c) a summer coupled to the input signal compensation block and to the power signal compensation block for providing an error signal corresponding to a difference between the target power level and the power level of the total power signal; (d) a transient detect block coupled to the input signal compensation block for providing a transient signal to identify a transient condition when the rate of change in the slew rate of the compensated input signal exceeds a selected transient threshold; (e) a first transient control circuit coupled to the transient detect block for providing first and second digital transient control signals, wherein the first transient control signal indicates the occurrence of a transient condition for a first time period in response to the transient signal and wherein the second transient control signal indicates the occurrence of a transient condition for a second time period in response to the transient signal, and wherein the second time period is longer than the first time period; (f) a control circuit coupled to the summer for receiving an amplified error signal for providing a first control signal in response to the amplified error signal; (g) a signal combining block for combining the first control signal and the first transient control signal to provide a main power supply control signal; (h) a selectively engageable second transient control circuit coupled to the first transient control signal for receiving the second transient control circuit and for temporarily increasing the magnitude of the error signal, wherein the second transient control circuit is engaged and disengaged in response to the second transient control signal, the second transient control circuit including a feedback amplifier coupled between the summer and the control circuit to provide the amplified error signal, the feedback amplifier being operative at all times; and (i) a main power supply for providing a main power signal at the first power terminal in response to the main power supply control signal; wherein the total power signal corresponds to the main power signal.
 35. The power supply circuit of claim 34 wherein the first transient control circuit includes a first one-shot circuit for generating the first transient control signal, wherein the first one-shot circuit is triggered by the transient signal indicating the occurrence of a transient condition.
 36. The power supply circuit of claim 35 wherein the first transient control circuit includes a second one-shot circuit for generating the second transient control signal, wherein the second one-shot is triggered by the transient signal indicating the occurrence of a transient condition.
 37. The power supply circuit of claim 36 wherein the second transient control circuit includes: (i) a fast release block for initially increasing the error signal rapidly when the second transient control circuit becomes engaged; and (ii) a slow release block for slowly reducing the increase in the error signal.
 38. The power supply circuit of claim 37 wherein the first time period is selected to be longer than the time required for the fast attack block to increase the magnitude of the error signal.
 39. The power supply circuit of claim 37 wherein the second transient control circuit comprises: (i) a first diode having its cathode coupled to the output of the second one-shot circuit and having its anode coupled to first node; (ii) a first capacitor coupled between the first node and ground; (iii) a first resistor and a second capacitor coupled in parallel between the first node and a second node, wherein the second node is coupled to a third node at the coupling of the summer and the power signal compensation block; and (iv) a feedback network including a second resistor and a third capacitor coupled between the third node and a fourth node at the coupling of the feedback amplifier and the control circuit.
 40. The power supply circuit of claim 39 wherein: (i) the second transient control signal is normally high in the absence of a transient condition and becomes low when the second one-shot is triggered; (ii) the first capacitor is normally charged in the absence of a transient condition and is discharged through the first diode when the second one-shot is triggered; and (iii) the first and second resistances act as a voltage divider in response to the discharging of the first capacitor to initially increase the magnitude of the amplified error signal.
 41. The power supply circuit of claim 37 wherein the signal combining block includes an OR gate.
 42. The power supply circuit of claim 39 wherein the first input compensation circuit includes: (i) an offset block for adding an offset to said input signal to provide an offset input signal; (ii) a frequency compensation block for receiving the offset input signal and for providing a corresponding frequency compensated circuit having its voltage components phase advanced with respect to its current component; and (iii) a first rectifier for rectifying the frequency compensated circuit to provide the compensated input signal.
 43. The power supply circuit of claim 42 wherein the frequency compensation block is configured to amplify the amplitude of selected the frequency compensated signal at selected frequency components, wherein said selected frequencies exceed a selected frequency compensation threshold.
 44. The power supply circuit of claim 43 wherein the amplitude of the selected frequency components is progressively amplified to a greater extent.
 45. The power supply circuit of claim 43 wherein the amplitude of the selected frequency components is equally amplified.
 46. The power supply circuit of claim 42 wherein the first rectifier is a half wave rectifier.
 47. The power supply circuit of claim 42 wherein the input offset block is configured to add a smaller offset to the first input signal if the highest frequency component of the input signal is less than a selected offset frequency threshold and to add a larger offset to the first input signal otherwise.
 48. The power supply circuit of claim 42 wherein the first input compensation block provides the compensated input signal corresponding to a target power level that exceeds the sum of the power required by the first amplifier to generate a first output signal corresponding to the first input signal and at least half of a ripple in the main power signal.
 49. The power supply circuit of claim 40 wherein the first control circuit is a PWM signal having a fixed switching frequency.
 50. The power supply circuit of claim 39 wherein the main power supply is a switching regulator including: (i) a main power source; (ii) a switch coupled to the power source and responsive to the first control signal to provide an unfiltered main power signal; and (iii) an integrating filter coupled to switch to provide the main power signal corresponding to the unfiltered power signal.
 51. The power supply circuit of claim 50 wherein the switching frequency is selected to limit the EMI emitted by the main power supply to a selected maximum EMI limit.
 52. The power supply of claim 50 wherein the transient detect block includes: (i) a peak detector for providing a peak signal corresponding to the peak envelope of the compensated input signal; (ii) a differentiator coupled to the peak detector for providing a differentiated signal corresponding to the rate of change of the compensated input signal; and (iii) a comparator for comparing the differentiated signal with the transient threshold to provide the transient signal.
 53. The power supply circuit of claim 51 wherein a time constant of the integrating filter is selected to effectively smooth the main power signal compared to the unfiltered power signal.
 54. The power supply circuit of claim 52 wherein a discharge rate of the peak detector is selected to correspond to a slew rate of the main power supply.
 55. The power supply circuit of claim 40 further including: (i) an overload detect block coupled to the first amplifier to provide an overload signal corresponding to one or more overload conditions within the first amplifier; and (ii) means for combining the overload signal with the compensated input signal to provide an adjusted compensated input signal; wherein the error signal corresponds to the adjusted compensated input signal and compensated power signal.
 56. The power supply circuit of claim 55 wherein the means for combining is the summer.
 57. The power supply circuit of claim 55 wherein the means for combining is a second summer.
 58. The power supply circuit of claim 40 wherein a second amplifier is coupled to the power terminal and a second input signal is received at a second input terminal and wherein the first input signal compensation block provides a first compensated input signal and further including: (i) a second input signal compensation from for providing a second compensated input signal; and (ii) a combining circuit for combining the first and second compensated input signals to provide the compensated input signal having a magnitude corresponding to the higher of the first and second compensated input signals.
 59. The power supply circuit of claim 58 wherein the combining circuit is a pair of diodes for diode OR'ing the first and second compensated input signals.
 60. The power supply circuit of claim 40 wherein the first control signal is a pulse density modulated control signal and wherein the main power supply is a resonant switching power regulator.
 61. The power supply circuit of claim 60 wherein the main power supply is a zero-current switching regulator and includes a LC resonant tank.
 62. The power supply circuit of claim 40 further including a low voltage power supply coupled for providing a fixed low voltage power signal to the power terminal, wherein the total power signal is generally equal to the higher of the magnitude of the main power signal, the magnitude of the transient power signal or the magnitude of the low voltage power signal.
 63. The power supply circuit of claim 62 wherein the control circuit is configured to set the main power signal to zero when the target power level is less than the magnitude of the low voltage power signal.
 64. The power supply circuit of claim 60 further including a post regulation circuit having: (i) an overload detect block coupled to the first amplifier to provide an overload signal corresponding to one or more overload conditions within the first amplifier; (ii) a second rectifier coupled to the offset block for receiving the offset input signal and providing a rectified input signal; (iii) a third summer for subtracting the overload signal from the rectified input signal to provide a regulation signal; (iv) a regulation amplifier coupled to the third summer for providing a amplified regulation signal corresponding to the regulation signal and having a magnitude range corresponding to the magnitude range of the total power signal; (v) a post regulator having a control terminal, and coupled between the main and transient power supplies and the power terminal; and (vi) a regulation feedback circuit coupled between the power terminal and the control terminal of the post regulator and including a fourth summer for providing a regulator error signal corresponding to the difference between the total power signal and the amplified regulation signal; wherein the post regulator regulates the total power signal in response to the regulator error signal when an overload condition occurs.
 65. The power supply circuit of any one of claims 40-45 or 47-64 wherein the first amplifier is a bridge amplifier and wherein the first rectifier is a full wave rectifier.
 66. The power amplifier of any of claims 40-64 wherein the first signal amplifier has a second power terminal and further including a second power supply circuit having the same structure as said first power supply circuit, wherein said first power supply circuit supplies power to said first signal amplifier at said first power terminal during positive half wave of said output signal and said second power supply circuit provides power to said first signal amplifier at said second power terminal during negative half waves of said output signal.
 67. The power amplifier of claim 34 further including an EMI isolation circuit coupled between said first input terminal and an internal input terminal for providing a first EMI-decoupled signal corresponding to said first input signal at said internal input terminal, and wherein said first input signal compensation block and said first signal amplifier are coupled to said internal input node, wherein the EMI isolation circuit has: (i) a first isolation amplifier having a first amplification factor coupled to said first input terminal through a first impedance for receiving said input signal and for providing an amplified input signal; and (ii) a second isolation amplifier having a second amplification factor coupled to said first amplifier for receiving said amplified input signal and to said internal input terminal for providing said first EMI-decoupled signal; wherein the amplification factor of the first isolation amplifier is greater than 1 and the amplification factor of said second isolation amplifier is less than
 1. 68. The power amplifier of claim 64 wherein said first impedance is greater than 50 kΩ.
 69. The power amplifier of claim 64 wherein said first impedance is equal to or greater than 100 kΩ.
 70. A method of supplying a total power signal to a signal amplifier, comprising: (a) receiving an input signal; (b) producing a compensated input signal corresponding to the input signal, the compensated input signal defining a target power level for the power signal; (c) comparing the compensated signal to a reduced version of the power signal to produce an error signal; (d) providing first and second control signals in response to the error signal; (e) providing a main power signal using a switching regulator in response to the first control signal, the main power signal being a first part of the total power signal; (f) comparing the rate of change of the compensated input signal to a selected transient threshold to provide a transient signal, the transient signal identifying a transient condition when the rate of change exceeds the threshold, the transient threshold corresponding to the maximum slew rate of the main power supply; and (g) engaging a transient power supply to provide a transient power signal in response to said second control signal, when the transient signal indicates a transient condition, the transient power signal being a second part of the total power signal.
 71. The method of claim 70 wherein step (b) includes: (i) adding an offset the input signal; (ii) amplifying frequency components of the input signal exceeding a selected threshold frequency; and (iii) rectifying the result of step (b).
 72. The method of claim 71 wherein step (i) is performed after step (iii).
 73. The method of claim 71 wherein step (i) is performed by adding a smaller offset to frequency components below a selected threshold and adding a larger offset to frequency components above the threshold.
 74. The method of claim 70 wherein step (f) is performed by: (i) peak detecting the compensated input signal; (ii) differentiating the result of step (i); and (iii) comparing the result of step (ii) with the transient threshold.
 75. The method of claim 74 wherein a discharge rate of the peak detector is selected to correspond to the slew rate of the main power signal.
 76. The method of claim 70 further including providing an overload signal corresponding to an overload condition in the amplifier and reducing the magnitude of the main power signal in response to the overload signal.
 77. The method of claim 70 wherein the first control signal is a PWM signal.
 78. The method of claim 70 wherein the switching regulator is a resonant switching power regulator, and wherein the first control signal is a PDM signal.
 79. The method of claim 70 further including providing a low voltage DC power signal as a third part of the total power signal.
 80. The method of claim 79 further including disabling the switching regulator when the target power level is less than the magnitude of the low voltage DC power signal.
 81. The method of claim 70 further including regulating the total power signal using a post regulator.
 82. The method of claim 81 further including providing an overload signal corresponding to an overload condition in the amplifier and reducing the magnitude of the total power signal in response to the overload signal.
 83. The method of claim 71 wherein the amplifier is a bridge amplifier and step (iii) is performed by full wave rectifying the result of step (ii).
 84. A method of supplying a total power signal to a signal amplifier, comprising: (a) receiving an input signal; (b) producing a compensated input signal corresponding to the input signal, the compensated input signal defining a target power level for the power signal; (c) comparing the compensated signal to a reduced version of the power signal to produce an error signal; (d) providing first and second control signals in response to the error signal; (e) providing a main power signal using a switching regulator in response to the first control signal, the main power signal being a first part of the total power signal; (f) comparing the rate of change of the compensated input signal to a selected transient threshold to provide a transient signal, the transient signal identifying a transient condition when the rate of change exceeds the threshold, the transient threshold corresponding to the maximum slew rate of the main power supply; and (g) in response to a transient condition, temporarily engaging the switching regulator with a 100% duty cycle for a first time period and temporarily elevating the error signal for a second time period.
 85. The method of claim 84 wherein step (b) includes: (i) adding an offset the input signal; (ii) amplifying frequency components of the input signal exceeding a selected threshold frequency; and (iii) rectifying the result of step (b).
 86. The method of claim 85 wherein step (i) is performed after step (iii).
 87. The method of claim 85 wherein step (i) is performed by adding a smaller offset to frequency components below a selected threshold and adding a larger offset to frequency components above the threshold.
 88. The method of claim 84 wherein step (f) is performed by: (i) peak detecting the compensated input signal; (ii) differentiating the result of step (i); and (iii) comparing the result of step (ii) with the transient threshold.
 89. The method of claim 88 wherein a discharge rate of the peak detector is selected to correspond to the slew rate of the main power signal.
 90. The method of claim 84 further including providing an overload signal corresponding to an overload condition in the amplifier and reducing the magnitude of the main power signal in response to the overload signal.
 91. The method of claim 84 wherein the first control signal is a PWM signal.
 92. The method of claim 84 wherein the switching regulator is a resonant switching power regulator, and wherein the first control signal is a PDM signal.
 93. The method of claim 84 further including providing a low voltage DC power signal as a second part of the total power signal.
 94. The method of claim 93 further including disabling the switching regulator when the target power level is less than the magnitude of the low voltage DC power signal.
 95. The method of claim 84 further including regulating the total power signal using a post regulator.
 96. The method of claim 95 further including providing an overload signal corresponding to an overload condition in the amplifier and reducing the magnitude of the total power signal in response to the overload signal.
 97. The method of claim 85 wherein the amplifier is a bridge amplifier and step (iii) is performed by full wave rectifying the result of step (ii).
 98. A power amplifier for receiving a first input signal at a first input terminal and for producing a first output signal at a first output terminal, said first output signal corresponding to said first input signal, a first signal amplifier being coupled to the input terminal to receive the input signal and coupled to the output terminal to provide the output signal, the first signal amplifier having a first power terminal for receiving a total power signal and said power amplifier having a first power supply circuit comprising: (a) a first input signal compensation block coupled to the first input terminal to receive the first input signal and to provide a compensated input signal corresponding to the first input signal, wherein the compensated input signal defines a target power level; (b) a main power signal compensation block for receiving a main power signal and for providing a compensated main power signal corresponding to the main power signal; (c) a first summer coupled to the input signal compensation block and to the main power signal compensation block for providing a first error signal corresponding to a difference between the target power level and the power level of the main power signal; (d) a first control circuit coupled to the summer for receiving the first error signal and for providing a first control signal in response to the second error signal, wherein the first control signal corresponds to a target main power signal level; (e) a total power signal compensation block for receiving the total power signal and for providing a compensated total power signal corresponding to the total power signal; (f) a second summer coupled to the input signal compensation block and to the total power signal compensation block for providing a second error signal corresponding to a difference between the target power level and the power level of the total power signal; (g) a second control circuit coupled to the summer for receiving the second error signal and for providing a second control signal in response to the second error signal, wherein the second control signal corresponds to a target transient power signal level; (h) a transient detect block coupled to the input signal compensation block for providing a transient signal to identify a transient condition when the rate of change in the slew rate of the compensated input signal exceeds a selected transient threshold; (i) a main power supply for providing a main power signal at the first power terminal in response to the first control signal; and (j) a selectively engageable transient power supply for providing a transient power signal at the first power terminal in response to the second control signal and the transient signal, wherein the transient power supply is engaged when the transient signal indicates that a transient condition exists; wherein the magnitude of the total power signal is generally equal to the higher of the magnitude of the main power signal or the magnitude of the transient power signal. 